ISL78229 Datasheet, PDF (65/71 Pages) Intersil Corporation – 2-Phase Boost Controller with Drivers

ISL78229
Application Information
There are several ways to define the external components and
parameters of boost regulators. This section shows one example
of how to decide the parameters of the external components
based on the typical application schematics as shown in Figure 4
on page 8. In the actual application, the parameters may need to
be adjusted and additional components may be needed for the
specific applications regarding noise, physical sizes, thermal,
testing and/or other requirements.
Output Voltage Setting
The output voltage (V_OUT) of the regulator can be programmed by
an external resistor divider connecting from V_OUT to FB and FB to
GND as shown in Figure 4 on page 8. Use Equation 2 on page 26
to calculate the desired V_OUT, where V_REF can be either
VREF_DAC or VREF_TRK, whichever is lower. VREF_DAC default is
1.6V and can be programmed to a value between 0V to 2.04V via
PMBus™ command “VOUT_COMMAND (21h)” on page 48. In the
actual application, the resistor value should be decided by
considering the quiescent current requirement and loop
response. Typically, between 4.7kΩ to 20kΩ will be used for the
R_FB1.
Switching Frequency
Switching frequency is determined by requirements of transient
response time, solution size, EMC/EMI, power dissipation and
efficiency, ripple noise level, input and output voltage range.
Higher frequency may improve the transient response and help
to reduce the solution size. However, this may increase the
switching losses and EMC/EMI concerns. Thus, a balance of
these parameters are needed when deciding the switching
frequency.
Once the switching frequency f_SW is decided, the frequency
setting resistor (R_FSYNC) can be determined by Equation 6 on
page 29.
Input Inductor Selection
While the boost converter is operating in steady state Continuous
Conduction Mode (CCM), the output voltage is determined by
Equation 1 on page 25. With the required input and output voltage,
duty cycle D can be calculated by Equation 34:
D = 1 – _V----^V-_O---^I-_U-^N---_T--
(EQ. 34)
Where D is the on-duty of the boost low-side power transistor.
Under this CCM condition, the inductor peak-to-peak ripple
current of each phase can be calculated as Equation 35:
I_L(P-P) = D  T  ^V----_L-^I--^N---
(EQ. 35)
Where T is the switching cycle 1/f_SW and L is each phase
inductor's inductance.
From the previous equations, the inductor value is determined by
Equation 36:
L
=



1
–
V^----V-_O^---I-_U^-N---_T^--_{}
 _I--_L---₍--_P----_--^V_P----_)I--^N-_---_f--_S----_W---
(EQ. 36)
Use Equation 36 to calculate L, where values of V_IN, V_OUT and I_Lpp
are based on the considerations described in the following:
• One method is to select the minimum input voltage and the
maximum output voltage under long term operation as the
conditions to select the inductor. In this case, the inductor DC
current is the largest.
• The general rule to select inductor is to have its ripple current
I_L(P-P) around 30% to 50% of maximum DC current. The
individual maximum DC inductor current for the 2-phase boost
converter can be calculated by Equation 37, where P_OUTmax is
the maximum DC output power, EFF is the estimated efficiency:
I_Lmax = _V-----_I--_N--^P--_m--^O---_i-_n-^U----_^T---_E^m----_F-^a---_F-^x----_---₂--
(EQ. 37)
Using Equation 36 with the two conditions listed above, a
reasonable starting point for the minimum inductor value can be
estimated from Equation 38, where K is typically selected as
30%.
L_min
=


1

–
V^----V-_O^----IU-N---_T^--m--_m^--i--n-_a^---_x^-_{}
 _P----^V-_O---^I2-_U-^N---_T-^m---_m--ⁱ-ⁿ--_a---^_x---^E-_---_K-^F----^F-_---_f-^-_S--²--_W---
(EQ. 38)
Increasing the value of the inductor reduces the ripple current
and therefore the ripple voltage. However, the large inductance
value may reduce the converter’s response time to a load
transient. Also, this reduces the ramp signal and may cause a
noise sensitivity issue.
The peak current at maximum load condition must be lower than
the saturation current rating of the inductor with enough margin.
In the actual design, the largest peak current may be observed at
some transient conditions like the start-up or heavy load
transient. Therefore, the inductor’s size needs to be determined
with the consideration of these conditions. To avoid exceeding
the inductor’s saturation rating, OC1 peak current limiting (refer
to “Peak Current Cycle-by-Cycle Limiting (OC1)” on page 36) should
be selected below the inductor’s saturation current rating.
Output Capacitor
To filter the inductor current ripples and to have sufficient transient
response, output capacitors are required. A combination of
electrolytic and ceramic capacitors are normally used.
The ceramic capacitors are used to filter the high frequency
spikes of the main switching devices. In layout, these output
ceramic capacitors must be placed as close as possible to the
main switching devices to maintain the smallest switching loop
in layout. To maintain capacitance over the biased voltage and
temperature range, good quality capacitors such as X7R or X5R
are recommended.
The electrolytic capacitors are normally used to handle the load
transient and output ripples. The boost output ripples are mainly
dominated by the load current and output capacitance volume.
For boost converter, the maximum output voltage ripple can be
estimated using Equation 39, where I_OUTmax is the load current
at output, C is the total capacitance at output, and D_MIN is the
minimum duty cycle at VIN_max and VOUT_min.
V_OUTripple= ^I--^O-----^U----^T----^m-_C---^a---_-^x--₂--^---_--¹-_f--_S--–--_W---^D----^M-----^I--^N-----^
(EQ. 39)
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FN8656.3
February 12, 2016