ISL78227 Datasheet, PDF (37/43 Pages) Intersil Corporation – 2-Phase Boost Controller with Integrated Drivers

ISL78227
Based on the junction to ambient thermal resistance R_JA of the
package, the maximum junction temperature should be kept below
+125°C. However, the power losses at the LDO need to be
considered, especially when the gate drivers are driving external
MOSFETs with large gate charges. At high V_IN, the LDO has
significant power dissipation that may raise the junction
temperature where the thermal shutdown occurs.
With an external PNP transistor as shown in Figure 63, the power
dissipation of the internal LDO can be moved from the ISL78227
to the external transistor. Choose R_S to be 68Ω so that the LDO
delivers about 10mA when the external transistor begins to turn
on. The external circuit increases the minimum input voltage to
approximately 6.5V.
VIN
R_S
VIN
ISL78227
PVCC
PVCC
FIGURE 63. SUPPLEMENTING LDO CURRENT
Application Information
There are several ways to define the external components and
parameters of boost regulators. This section shows one example
of how to decide the parameters of the external components
based on the typical application schematics as shown in Figure 4
on page 8. In the actual application, the parameters may need to
be adjusted and additional components may be needed for the
specific applications regarding noise, physical sizes, thermal,
testing and/or other requirements.
Output Voltage Setting
The Output Voltage (V_OUT) of the regulator can be programmed
by an external resistor divider connecting from V_OUT to FB and FB
to GND as shown in Figure 4 on page 8. Use Equation 2 on
page 25 to calculate the desired V_OUT, where V_REF can be either
VREF_1.6V or VREF_TRK, whichever is lower. In the actual
application, the resistor value should be decided by considering
the quiescent current requirement and loop response. Typically,
between 4.7kΩ to 20kΩ will be used for the R_FB1.
Switching Frequency
Switching frequency is determined by requirements of transient
response time, solution size, EMC/EMI, power dissipation and
efficiency, ripple noise level, input and output voltage range.
Higher frequency may improve the transient response and help
to reduce the solution size. However, this may increase the
switching losses and EMC/EMI concerns. Thus, a balance of
these parameters are needed when deciding the switching
frequency.
Once the switching frequency f_SW is decided, the frequency
setting resistor (R_FSYNC) can be determined by Equation 6 on
page 28.
Input Inductor Selection
While the boost converter is operating in steady state Continuous
Conduction Mode (CCM), the output voltage is determined by
Equation 1 on page 24. With the required input and output voltage,
duty cycle D can be calculated by Equation 25:
D = 1 – _V----^V-_O---^I-_U-^N---_T--
(EQ. 25)
Where D is the on-duty of the boost low-side power transistor.
Under this CCM condition, the inductor peak-to-peak ripple
current of each phase can be calculated as Equation 26:
I_{LP-P} = D  T  ^V----_L-^I--^N---
(EQ. 26)
Where T is the switching cycle 1/f_SW and L is each phase
inductor’s inductance.
From the previous equations, the inductor value is determined by
Equation 27:
L
=



1
–
V^----V-_O^---I-_U^-N---_T^--_{}
 _I--_L----_--_P---_--^V-_P----^I_-^N--_---_f--_S----_W---
(EQ. 27)
Use Equation 27 to calculate L, where values of V_IN, V_OUT and
I_L(P-P) are based on the considerations described in following:
• One method is to select the minimum input voltage and the
maximum output voltage under long term operation as the
conditions to select the inductor. In this case, the inductor DC
current is the largest.
• The general rule to select inductor is to have its ripple current
I_L(P-P) around 30% to 50% of maximum DC current. The
individual maximum DC inductor current for the 2-phase boost
converter can be calculated by Equation 28, where P_OUTmax is
the maximum DC output power, EFF is the estimated efficiency:
I_Lmax = _V-----_I--_N--^P--_m--^O---_i-_n-^U----_^T---_E^m----_F-^a---_F-^x----_---₂--
(EQ. 28)
Using Equation 27 with the two conditions listed above, a
reasonable starting point for the minimum inductor value can be
estimated from Equation 29, where K is typically selected as
30%.
L_min
=


1

–
V^----V-_O^----IU-N---_T^--m--_m^--i--n-_a^---_x^-_{}
 _P----^V-_O---^I2-_U-^N---_T-^m---_m--ⁱ-ⁿ--_a---^_x---^E-_---_K-^F----^F-_---_f-^-_S--²--_W---
(EQ. 29)
Increasing the value of the inductor reduces the ripple current
and therefore the ripple voltage. However, the large inductance
value may reduce the converter’s response time to a load
transient. This also reduces the current sense ramp signal and
may cause a noise sensitivity issue.
The peak current at maximum load condition must be lower than
the saturation current rating of the inductor with enough margin.
In the actual design, the largest peak current may be observed at
some transient conditions like the start-up or heavy load
transient. Therefore, the inductor’s size needs to be determined
with the consideration of these conditions. To avoid exceeding
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FN8808.2
February 24, 2016