LMH2832 Datasheet, PDF (30/46 Pages) Texas Instruments – LMH2832 Fully Differential, Dual, 1.1-GHz, Digital Variable-Gain Amplifier

LMH2832
SBOS709A – JULY 2016 – REVISED JULY 2016
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Application Information (continued)
10.1.1.1 SNR Considerations
When using the LMH2832 with a filter, the signal-to-noise ratio (SNR) of the amplifier and filter can be calculated
from the amplitude of the signal and the bandwidth of the filter. The noise from the amplifier is band-limited by
the filter with the equivalent brick-wall filter bandwidth. The amplifier and filter noise can be calculated using
Equation 3:
× SNR_AMP+FILTER = 10 log
V^2O
= 20 × log
e^2FILTEROUT
V_O
e_FILTEROUT
where:
• e_FILTEROUT = e_NAMPOUT • √ENB
• e_NAMPOUT = the output noise density of the LMH2832 (50.4 nV/√Hz) at A_V = 30 dB
• ENB = the brick-wall equivalent noise bandwidth of the filter
• V_O = the amplifier output signal
(3)
For example, with a first-order (N = 1) band-pass or low-pass filter with a 1000-MHz cutoff, ENB is 1.57 • f_–3dB =
1.57 • 1000 MHz = 1570 MHz. For second-order (N = 2) filters, ENB is 1.22 • f_–3dB. When the filter order
increases, ENB approaches f_–3dB (N = 3 → ENB = 1.15 • f_–3dB; N = 4 → ENB = 1.13 • f_–3dB). Both V_O and
e_FILTEROUT are in RMS voltages. For example, with a 2-V_PP (0.707 V_RMS) output signal and a 300-MHz, first-order,
low-pass filter, the SNR of the amplifier and filter is 56 dB with e_FILTEROUT = 50.4 nV/√Hz • √471 MHz = 1.09
mV_RMS.
The SNR of the amplifier, filter, and ADC sum in RMS fashion, as shown in Equation 4 (SNR values in dB):
× SNR_SYSTEM = -20 log
-SNR_AMP+FILTER
-SNR_ADC
10 ¹⁰
+ 10 ¹⁰
(4)
This formula shows that if the SNR of the amplifier and filter equals the SNR of the ADC, then the combined
SNR is 3 dB lower (worse). Thus, for minimal degradation (< 1 dB) on the ADC SNR, the SNR of the amplifier
and filter must be 10 dB greater than the ADC SNR. The combined SNR calculated in this manner is usually
accurate to within ±1 dB of the actual implementation.
10.1.1.2 SFDR Considerations
The SFDR of the amplifier is usually set by the second- or third-order harmonic distortion for single-tone inputs,
and by the second-order or third-order intermodulation distortion for two-tone inputs. Harmonics and second-
order intermodulation distortion can be filtered to some degree, but third-order intermodulation spurs cannot be
filtered. The ADC generates the same distortion products as the amplifier, but also generates additional spurs
(not harmonically related to the input signal) as a result of sampling and clock feed through.
When the spurs from the amplifier and filter are known, each individual spur can be directly added to the same
spur from the ADC, as shown in Equation 5, to estimate the combined spur (spur amplitudes in dBc):
-HDx_AMP+FILTER
-HDx_ADC
× HDx_SYSTEM = -20 log 10 ²⁰
+ 10 ²⁰
(5)
This calculation assumes that the spurs are in phase, but usually provides a good estimate of the final combined
distortion.
For example, if the spur of the amplifier and filter equals the spur of the ADC, then the combined spur is 6 dB
higher. To minimize the amplifier contribution (< 1 dB) to the overall system distortion, the spur from the amplifier
and filter must be approximately 15 dB lower in amplitude than that of the converter. The combined spur
calculated in this manner is usually accurate to within ±6 dB of the actual implementation; however, higher
variations can be detected as a result of phase shift in the filter, especially in second-order harmonic
performance.
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