MAX1540A Datasheet, PDF (36/49 Pages) Maxim Integrated Products – Dual Step-Down Controllers with Saturation Protection, Dynamic Output, and Linear Regulator

Dual Step-Down Controllers with Saturation
Protection, Dynamic Output, and Linear Regulator
Inductor Selection
The switching frequency and inductor operating point
determine the inductor value as follows:
L=
( ) ^V_OUT ^V_IN ^{- V}_OUT
V_IN × f_SW x I_LOAD(MAX) × LIR
Setting the Current Limit
The minimum current-limit threshold must be great
enough to support the maximum load current when the
current limit is at the minimum tolerance value. The val-
ley of the inductor current occurs at I_LOAD(MAX) minus
half the ripple current; therefore:
For example: I_LOAD(MAX) = 4A, V_IN = 12V, V_OUT2 =
2.5V, f_SW = 355kHz, 30% ripple current or LIR = 0.3:
L = ^{2.5V × (12V - 2.5V)} = 4.65μH
12V × 355kHz x4A × 0.3
Find a low-loss inductor with the lowest possible DC
resistance that fits in the allotted dimensions. Ferrite
cores are often the best choice, although powdered
iron is inexpensive and can work well at 200kHz. The
core must be large enough not to saturate at the peak
inductor current (I_PEAK):
I_PEAK
=I_LOAD(MAX)
⎛⎝⎜₁₊
LIR ⎞
2 ⎠⎟
Most inductor manufacturers provide inductors in stan-
dard values, such as 1.0µH, 1.5µH, 2.2µH, 3.3µH, etc.
Also look for nonstandard values, which can provide a
better compromise in LIR across the input voltage
range. If using a swinging inductor (where the no-load
inductance decreases linearly with increasing current),
evaluate the LIR with properly scaled inductance values.
Transient Response
The inductor ripple current also impacts transient-
response performance, especially at low V_IN - V_OUT dif-
ferentials. Low inductor values allow the inductor
current to slew faster, replenishing charge removed
from the output filter capacitors by a sudden load step.
The amount of output sag is also a function of the maxi-
mum duty factor, which can be calculated from the on-
time and minimum off-time:
( ) ^VSAG
=
L(ΔI_LOAD(MAX)
)²
⎡⎛
⎣⎢⎢⎝⎜
V_OUT ×
V_IN
K
⎞
⎠⎟
+
t_OFF(MIN)
⎤
⎥
⎦⎥
⎡⎛
2C_OUT × V_{OUT ⎣⎢⎢⎝⎜}
V_IN - V_OUT
V_IN
×
K
⎞
⎟
⎠
-
t_OFF(MIN)
⎤
⎥
⎦⎥
where t_OFF(MIN) is the minimum off-time (see the
Electrical Characteristics) and K is from Table 3.
The amount of overshoot during a full-load to no-load tran-
sient due to stored inductor energy can be calculated as:
( ) _{VSOAR ≈}
ΔI_LOAD(MAX)
2
L
2C_OUT × V_OUT
I_LIM(VAL)
>
I_LOAD(MAX)
-
⎛
⎜
⎝
V_OUT(V_IN(MIN) - V_OUT
2V_IN(MIN) f_SW L
)
⎞
⎟
⎠
where I_LIM(VAL) equals the minimum valley current-limit
threshold voltage divided by the current-sense resis-
tance (R_SENSE). For the 50mV default setting, the mini-
mum valley current-limit threshold is 40mV.
Connect ILIM_ to V_CC for a default 50mV valley current-
limit threshold. In adjustable mode, the valley current-
limit threshold is precisely 1/10th the voltage seen at
ILIM_. For an adjustable threshold, connect a resistive
divider from REF to analog ground (GND) with ILIM_
connected to the center tap. The external 250mV to 2V
adjustment range corresponds to a 25mV to 200mV
valley current-limit threshold. When adjusting the
current limit, use 1% tolerance resistors and a divider
current of approximately 10µA to prevent significant
inaccuracy in the valley current-limit tolerance.
The current-sense method (Figure 14) and magnitude
determine the achievable current-limit accuracy and
power loss (Table 9). Typically, higher current-sense
voltage limits provide tighter accuracy, but also dissi-
pate more power. Most applications employ a valley
current-sense voltage (V_LIM(VAL)) of 50mV to 100mV,
so the sense resistor may be determined by:
R_SENSE = V_LIM(VAL) / I_LIM(VAL)
For the best current-sense accuracy and overcurrent
protection, use a 1% tolerance current-sense resistor
between the inductor and output as shown in Figure
14a. This configuration constantly monitors the inductor
current, allowing accurate valley current-limiting and
inductor-saturation protection.
For low-output-voltage applications that require higher
efficiency, the current-sense resistor can be connected
between the source of the low-side MOSFET (N_{L_}) and
power ground (Figure 14b) with CSN_ connected to the
drain of N_{L_} and CSP_ connected to power ground. In
this configuration, the additional current-sense resis-
tance only dissipates power when N_{L_} is conducting
current. Inductor-saturation protection must be dis-
abled with this configuration (LSAT = GND) since the
inductor current is only properly sensed when the low-
side MOSFET is turned on.
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