MAX1544 Datasheet, PDF (35/42 Pages) Maxim Integrated Products – Dual-Phase, Quick-PWM Controller for AMD Hammer CPU Core Power Supplies

Dual-Phase, Quick-PWM Controller for
AMD Hammer CPU Core Power Supplies
medium-sized MOSFETs. However, high-current appli-
cations driving large, high-side, MOSFETs require
boost capacitors larger than 0.1µF. For these applica-
tions, select the boost capacitors to avoid discharging
the capacitor more than 200mV while charging the
high-side MOSFET’s gates:
C_BST
=
N x Q_GATE
200mV
where N is the number of high-side MOSFETs used for
one regulator, and Q_GATE is the gate charge specified in
the MOSFET’s data sheet. For example, assume two
IRF7811W N-channel MOSFETs are used on the high
side. According to the manufacturer’s data sheet, a
single IRF7811W has a maximum gate charge of 24nC
(V_GS = 5V). Using the above equation, the required boost
capacitance would be:
C_BST
=
2 x 24nC
200mV
=
0.24µF
Selecting the closest standard value, this example
requires a 0.22µF ceramic capacitor.
Current-Balance Compensation (CCI)
The current-balance compensation capacitor (C_CCI)
integrates the difference between the main and sec-
ondary current-sense voltages. The internal compensa-
tion resistor (R_CCI = 20kΩ) improves transient response
by increasing the phase margin. This allows the
dynamics of the current balance loop to be optimized.
Excessively large capacitor values increase the inte-
gration time constant, resulting in larger current differ-
ences between the phases during transients.
Excessively small capacitor values allow the current
loop to respond cycle-by-cycle but can result in small
DC current variations between the phases. Likewise,
excessively large resistor values can also cause DC
current variations between the phases. Small resistor
values reduce the phase margin, resulting in marginal
stability in the current-balance loop. For most applica-
tions, a 470pF capacitor from CCI to the switching reg-
ulator’s output works well.
Connecting the compensation network to the output
(V_OUT) allows the controller to feed forward the output
voltage signal, especially during transients. To reduce
noise pickup in applications that have a widely distrib-
uted layout, it is sometimes helpful to connect the com-
pensation network to the quiet analog ground rather
than V_OUT.
Setting Voltage Positioning
Voltage positioning dynamically lowers the output volt-
age in response to the load current, reducing the
processor’s power dissipation. When the output is
loaded, an operational amplifier (Figure 5) increases
the signal fed back to the Quick-PWM controller’s feed-
back input. The adjustable amplification allows the use
of standard, current-sense resistor values, and signifi-
cantly reduces the power dissipated since smaller cur-
rent-sense resistors can be used. The load transient
response of this control loop is extremely fast, yet well
controlled, so the amount of voltage change can be
accurately confined within the limits stipulated in the
microprocessor power-supply guidelines.
The voltage-positioned circuit determines the load current
from the voltage across the current-sense resistors
(R_SENSE = R_CM = R_CS) connected between the inductors
and output capacitors, as shown in Figure 10. The voltage
drop can be determined by the following equation:
V_VPS = A_VPSI_LOADR_SENSE
A_VPS
=
η_SUM R_F
η_TOTAL R_B
where η_SUM is the number of phases summed together
for voltage-positioning feedback, and η_TOTAL is the total
number of active phases. When the slave controller is
disabled, the current-sense summation maintains the
proper voltage-positioned slope. Select the positive
input summing resistors so R_FBS = R_F and R_A = R_B.
Minimum Input Voltage Requirements
and Dropout Performance
The nonadjustable minimum off-time one-shot and the
number of phases restrict the output voltage adjustable
range for continuous-conduction operation. For best
dropout performance, use the slower (200kHz) on-time
settings. When working with low input voltages, the
duty-factor limit must be calculated using worst-case
values for on- and off-times. Manufacturing tolerances
and internal propagation delays introduce an error to
the TON K factor. This error is greater at higher fre-
quencies (Table 6). Also, keep in mind that transient
response performance of buck regulators operated too
close to dropout is poor, and bulk output capacitance
must often be added (see the V_SAG equation in the
Design Procedure section).
The absolute point of dropout is when the inductor cur-
rent ramps down during the minimum off-time (∆I_DOWN)
as much as it ramps up during the on-time (∆I_UP). The
ratio h = ∆I_UP/∆I_DOWN is an indicator of the ability to
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