MAX1519 Datasheet, PDF(34/43 Page) Maxim Integrated Products – Dual-Phase, Quick-PWM Controllers for Programmable CPU Core Power Supplies

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MAX1519 Datasheet, PDF (34/43 Pages) Maxim Integrated Products – Dual-Phase, Quick-PWM Controllers for Programmable CPU Core Power Supplies

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Dual-Phase, Quick-PWM Controllers for

Programmable CPU Core Power Supplies

Input Capacitor Selection

The input capacitor must meet the ripple current

requirement (IRMS) imposed by the switching currents.

The multiphase Quick-PWM controllers operate out-of-

phase, while the Quick-PWM slave controllers provide

selectable out-of-phase or in-phase on-time triggering.

Out-of-phase operation reduces the RMS input current

by dividing the input current between several stag-

gered stages. For duty cycles less than 100%/Î·OUTPH

per phase, the IRMS requirements may be determined

by the following equation:

IRMS

ï£«

ï£¬

ï£

Î·

ILOAD

OUTPH VIN

ï£¶

ï£·

ï£¸

Î·OUTPH VOUT(VIN â Î·OUTPHVOUT)

where Î·OUTPH is the total number of out-of-phase switch-

ing regulators. The worst-case RMS current requirement

occurs when operating with VIN = 2Î·OUTPHVOUT. At this

point, the above equation simplifies to IRMS = 0.5 Ã

ILOAD/Î·OUTPH.

For most applications, nontantalum chemistries (ceramic,

aluminum, or OS-CONâ¢) are preferred due to their resis-

tance to inrush surge currents typical of systems with a

mechanical switch or connector in series with the input. If

the Quick-PWM controller is operated as the second

stage of a two-stage power-conversion system, tantalum

input capacitors are acceptable. In either configuration,

choose an input capacitor that exhibits less than 10Â°C

temperature rise at the RMS input current for optimal cir-

cuit longevity.

Power MOSFET Selection

Most of the following MOSFET guidelines focus on the

challenge of obtaining high load-current capability

when using high-voltage (>20V) AC adapters. Low-cur-

rent applications usually require less attention.

The high-side MOSFET (NH) must be able to dissipate

the resistive losses plus the switching losses at both

VIN(MIN) and VIN(MAX). Calculate both of these sums.

Ideally, the losses at VIN(MIN) should be roughly equal to

losses at VIN(MAX), with lower losses in between. If the

losses at VIN(MIN) are significantly higher than the losses

at VIN(MAX), consider increasing the size of NH (reducing

RDS(ON) but with higher CGATE). Conversely, if the losses

at VIN(MAX) are significantly higher than the losses at

VIN(MIN), consider reducing the size of NH (increasing

RDS(ON) to lower CGATE). If VIN does not vary over a wide

range, the minimum power dissipation occurs where the

resistive losses equal the switching losses.

Choose a low-side MOSFET that has the lowest possi-

ble on-resistance (RDS(ON)), comes in a moderate-

sized package (i.e., one or two SO-8s, DPAK, or

D2PAK), and is reasonably priced. Ensure that the DL

gate driver can supply sufficient current to support the

gate charge and the current injected into the parasitic

gate-to-drain capacitor caused by the high-side MOSFET

turning on; otherwise, cross-conduction problems can

occur (see the MOSFET Gate Driver section).

MOSFET Power Dissipation

Worst-case conduction losses occur at the duty factor

extremes. For the high-side MOSFET (NH), the worst-

case power dissipation due to resistance occurs at the

minimum input voltage:

(NH

RESISTIVE)

ï£«

ï£ï£¬

VOUT

VIN

ï£¶

ï£¸ï£·

ï£«

ï£¬

ï£

ILOAD

Î· TOTAL

ï£¶2

ï£·

ï£¸

RDS(ON)

where Î·TOTAL is the total number of phases.

Generally, a small high-side MOSFET is desired to

reduce switching losses at high input voltages.

However, the RDS(ON) required to stay within package

power dissipation often limits how small the MOSFETs

can be. Again, the optimum occurs when the switching

losses equal the conduction (RDS(ON)) losses. High-

side switching losses do not usually become an issue

until the input is greater than approximately 15V.

Calculating the power dissipation in high-side

MOSFETs (NH) due to switching losses is difficult since

it must allow for difficult quantifying factors that influ-

ence the turn-on and turn-off times. These factors

include the internal gate resistance, gate charge,

threshold voltage, source inductance, and PC board

layout characteristics. The following switching-loss cal-

culation provides only a very rough estimate and is no

substitute for breadboard evaluation, preferably includ-

ing verification using a thermocouple mounted on NH:

(NH

SWITCHING)

(VIN(MAX))2

ï£«

ï£ï£¬

CRSSfSW

IGATE

ï£¶

ï£¸ï£·

ï£«

ï£¬

ï£

ILOAD

Î· TOTAL

ï£¶

ï£·

ï£¸

where CRSS is the reverse transfer capacitance of NH and

IGATE is the peak gate-drive source/sink current (1A, typ).

Switching losses in the high-side MOSFET can become

an insidious heat problem when maximum AC adapter

voltages are applied, due to the squared term in the C

Ã VIN2 Ã fSW switching-loss equation. If the high-side

MOSFET chosen for adequate RDS(ON) at low battery

voltages becomes extraordinarily hot when biased from

VIN(MAX), consider choosing another MOSFET with

lower parasitic capacitance.

OS-CON is a trademark of Sanyo.

34 ______________________________________________________________________________________

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