LM3533 Datasheet, PDF (48/53 Pages) Texas Instruments – Complete Lighting Power Solution for Smartphone Handsets

LM3533
tolerance and DC bias response. For proper operation the degradation in capacitance due to tolerance, DC bias, and temperature,
should stay above 0.4µF. This might require placing two devices in parallel in order to maintain the required output capacitance
over the device operating range, and series LED configuration.
Because the output capacitor is in the path of the inductor current discharge path it will see a high-current step from 0 to I_PEAK each
time the switch turns off and the Schottky diode turns on. Any inductance along this series path from the cathode of the diode
through COUT and back into the LM3533's GND pin will contribute to voltage spikes (V_SPIKE = LP_ × dI/dt) at SW and OUT. These
spikes can potentially over-voltage the SW pin, or feed through to GND. To avoid this, COUT+ must be connected as close as
possible to the Cathode of the Schottky diode, and COUT− must be connected as close as possible to the LM3533's GND bump.
The best placement for COUT is on the same layer as the LM3533 so as to avoid any vias that can add excessive series inductance.
Schottky Diode Placement
The Schottky diode must have a reverse breakdown voltage greater than the LM3533’s maximum output voltage (see OVER-
VOLTAGE PROTECTION (INDUCTIVE BOOST) section). Additionally, the diode must have an average current rating high enough
to handle the LM3533’s maximum output current, and at the same time the diode's peak current rating must be high enough to
handle the peak inductor current. Schottky diodes are required due to their lower forward voltage drop (0.3V to 0.5V) and their fast
recovery time.
In the LM3533’s boost circuit the Schottky diode is in the path of the inductor current discharge. As a result the Schottky diode sees
a high-current step from 0 to I_PEAK each time the switch turns off and the diode turns on. Any inductance in series with the diode
will cause a voltage spike (V_SPIKE = LP_ × dI/dt) at SW and OUT. This can potentially over-voltage the SW pin, or feed through to
VOUT and through the output capacitor and into GND. Connecting the anode of the diode as close as possible to the SW pin and
the cathode of the diode as close as possible to COUT+ will reduce the inductance (LP_) and minimize these voltage spikes.
Inductor Placement
The node where the inductor connects to the LM3533’s SW bump has 2 issues. First, a large switched voltage (0 to VOUT +
VF_SCHOTTKY) appears on this node every switching cycle. This switched voltage can be capacitively coupled into nearby nodes.
Second, there is a relatively large current (input current) on the traces connecting the input supply to the inductor and connecting
the inductor to the SW bump. Any resistance in this path can cause voltage drops that can negatively affect efficiency and reduce
the input operating voltage range.
To reduce the capacitive coupling of the signal on SW into nearby traces, the SW bump-to-inductor connection must be minimized
in area. This limits the PCB capacitance from SW to other traces. Additionally, high-impedance nodes that are more susceptible
to electric field coupling need to be routed away from SW and not directly adjacent or beneath. This is especially true for traces
such as SCL, SDA, HWEN, PWM, and possibly ALS. A GND plane placed directly below SW will dramatically reduce the capaci-
tance from SW into nearby traces.
Lastly, limit the trace resistance of the VBATT-to-inductor connection and from the inductor-to-SW connection, by use of short,
wide traces.
Boost Input Capacitor Selection and Placement
The input capacitor on the LM3533 filters the voltage ripple due to the switching action of the inductive boost and the capacitive
charge pump doubler. A ceramic capacitor of at least 2.2µF must be used.
For the LM3533’s boost converter, the input capacitor filters the inductor current ripple and the internal MOSFET driver currents
during turn on of the internal power switch. The driver current requirement can range from 50mA at 2.7V to over 200mA at 5.5V
with fast durations of approximately 10ns to 20ns. This will appear as high di/dt current pulses coming from the input capacitor
each time the switch turns on. Close placement of the input capacitor to the IN pin and to the GND pin is critical since any series
inductance between IN and CIN+ or CIN− and GND can create voltage spikes that could appear on the VIN supply line and in the
GND plane.
Close placement of the input bypass capacitor at the input side of the inductor is also critical. The source impedance (inductance
and resistance) from the input supply, along with the input capacitor of the LM3533, form a series RLC circuit. If the output resistance
from the source (RS) is low enough the circuit will be underdamped and will have a resonant frequency (typically the case). De-
pending on the size of LS the resonant frequency could occur below, close to, or above the LM3533's switching frequency. This
can cause the supply current ripple to be:
1. Approximately equal to the inductor current ripple when the resonant frequency occurs well above the LM3533's switching
frequency;
2. Greater than the inductor current ripple when the resonant frequency occurs near the switching frequency; or
3. Less than the inductor current ripple when the resonant frequency occurs well below the switching frequency. Figure 19 shows
the series RLC circuit formed from the output impedance of the supply and the input capacitor.
The circuit is redrawn for the AC case where the VIN supply is replaced with a short to GND and the LM3533 + Inductor is replaced
with a current source (ΔIL). Equation 1 is the criteria for an underdamped response. Equation 2 is the resonant frequency. Equation
3 is the approximated supply current ripple as a function of LS, RS, and CIN.
As an example, consider a 3.6V supply with 0.1Ω of series resistance connected to C_IN through 50nH of connecting traces. This
results in an under-damped input-filter circuit with a resonant frequency of 712kHz. Since both the 1MHz and 500kHz switching
frequency options lie close to the resonant frequency of the input filter, the supply current ripple is probably larger than the inductor
current ripple. In this case, using equation 3, the supply current ripple can be approximated as 1.68 times the inductor current ripple
(using a 500kHz switching frequency) and 0.86 times the inductor current ripple using a 1MHz switching frequency. Increasing the
series inductance (LS) to 500nH causes the resonant frequency to move to around 225kHz, and the supply current ripple to be
approximately 0.25 times the inductor current ripple (500kHz switching frequency) and 0.053 times for a 1MHz switching frequency.
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