THS6032 Datasheet, PDF (19/31 Pages) Texas Instruments – LOW-POWER ADSL CENTRAL-OFFICE LINE DRIVER

THS6032
LOW-POWER ADSL CENTRAL-OFFICE LINE DRIVER
SLOS233C – APRIL1999 – REVISED MARCH 2000
APPLICATION INFORMATION
PCB design considerations (continued)
D Proper power supply decoupling – Use a minimum of a 6.8-µF tantalum capacitor in parallel with a 0.1-µF
ceramic capacitor on each supply terminal. It may be possible to share the tantalum among several
amplifiers depending on the application, but a 0.1-µF ceramic capacitor should always be used on the
supply terminal of every amplifier. In addition, the 0.1-µF capacitor should be placed as close as possible
to the supply terminal. As this distance increases, the inductance in the connecting etch makes the capacitor
less effective. The designer should strive for distances of less than 0.1 inches between the device power
terminal and the ceramic capacitors.
D Differential power supply decoupling – The THS6032 was designed for driving low-impedance differential
signals. The 25 Ω load which each amplifier drives causes large amounts of currents to flow from amplifier
to amplifier. Power supply decoupling for differential current signals must be accounted for to ensure low
distortion of the THS6032. By simply connecting a 0.1-µF ceramic capacitor from the +V_CC(H) pin to the
–V_CC(H) pin, along with another 0.1-µF ceramic capacitor from the +V_CC(L) pin to the –V_CC(L) pin, differential
current loops will be minimized (see Figure 36). This will help keep the THS6032 operating at peak
performance.
recommended feedback and gain resistor values
As with all current feedback amplifiers, the bandwidth of the THS6032 is an inversely proportional function of
the value of the feedback resistor. This can be seen from Figures 1 to 6. The recommended resistors for the
optimum frequency response with a 25-Ω load system can be seen in Table 1. These should be used as a
starting point and once optimum values are found, 1% tolerance resistors should be used to maintain frequency
response characteristics. For most applications, a feedback resistor value of 1.3 kΩ is recommended, which
is a good compromise between bandwidth and phase margin that yields a very stable amplifier.
Consistent with current feedback amplifiers, increasing the gain is best accomplished by changing the gain
resistor, not the feedback resistor. This is because the bandwidth of the amplifier is dominated by the feedback
resistor value and the internal dominant-pole capacitor. The ability to control the amplifier gain independently
of the bandwidth constitutes a major advantage of current feedback amplifiers over conventional voltage
feedback amplifiers. Therefore, once a frequency response is found suitable to a particular application, adjust
the value of the gain resistor to increase or decrease the overall amplifier gain.
Finally, it is important to realize the effects of the feedback resistance on distortion. Increasing the resistance
decreases the loop gain and increases the distortion. It is also important to know that decreasing load
impedance increases total harmonic distortion (THD). Typically, the third order harmonic distortion increases
more than the second order harmonic distortion.
Table 1. Recommended Feedback Resistor Values for 25 Ω Loads
GAIN
1
2, – 1
5
7.8
10
Rf
1.3 kΩ
1.1 kΩ
820 Ω
680 Ω
510 Ω
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