LM3481 Datasheet, PDF (9/22 Pages) National Semiconductor (TI) – High Efficiency Low-Side N-Channel Controller for Switching Regulators

FIGURE 1. Basic Operation of the PWM comparator
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OVER VOLTAGE PROTECTION
The LM3481 has over voltage protection (OVP) for the output
voltage. OVP is sensed at the feedback pin (FB). If at anytime
the voltage at the feedback pin rises to V_FB + V_OVP, OVP is
triggered. See the electrical characteristics section for limits
on V_FB and V_OVP.
OVP will cause the drive pin (DR) to go low, forcing the power
MOSFET off. With the MOSFET off, the output voltage will
drop. The LM3481 will begin switching again when the feed-
back voltage reaches V_FB + (V_OVP - V_OVP(HYS)). See the elec-
trical characteristics section for limits on V_OVP(HYS).
The internal bias of the LM3481 comes from either the internal
bias voltage generator as shown in the block diagram or di-
rectly from the voltage at the VIN pin. At input voltages lower
than 6V the internal IC bias is the input voltage and at voltages
above 6V the internal bias voltage generator of the LM3481
provides the bias.
SLOPE COMPENSATION RAMP
The LM3481 uses a current mode control scheme. The main
advantages of current mode control are inherent cycle-by-cy-
cle current limit for the switch and simpler control loop char-
acteristics. It is easy to parallel power stages using current
mode control since current sharing is automatic. However
there is a natural instability that will occur for duty cycles, D,
greater than 50% if additional slope compensation is not ad-
dressed as described below.
The current mode control scheme samples the inductor cur-
rent, I_L, and compares the sampled signal, V_samp, to a inter-
nally generated control signal, V_c. The current sense resistor,
R_SEN, as shown in Figure 5, converts the sampled inductor
current, I_L, to the voltage signal, V_samp, that is proportional to
I_L such that:
V_samp = I_L x R_SEN
The rising and falling slopes, M₁ and −M₂ respectively, of
V_samp are also proportional to the inductor current rising and
falling slopes, M_on and −M_off respectively. Where M_on is the
inductor slope during the switch on-time and −M_off is the in-
ductor slope during the switch off-time and are related to M₁
and −M₂ by:
M₁ = M_on x R_SEN
−M₂ = −M_off x R_SEN
For the boost topology:
M_on = V_IN / L
−M_off = (V_IN − V_OUT) / L
M₁ = [V_IN / L] x R_SEN
−M₂ = [(V_IN − V_OUT) / L] x R_SEN
M₂ = [(V_OUT − V_IN) / L] x R_SEN
Current mode control has an inherent instability for duty cy-
cles greater than 50%, as shown in Figure 2, where the control
signal slope, M_C, equals zero. In Figure 2, a small increase in
the load current causes the sampled signal to increase by
ΔV_samp0. The effect of this load change, ΔV_samp1, at the end
of the first switching cycle is :
From the above equation, when D > 0.5, ΔV_samp1 will be
greater than ΔV_samp0. In other words, the disturbance is di-
vergent. So a very small perturbation in the load will cause
the disturbance to increase. To ensure that the perturbed sig-
nal converges we must maintain:
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