OPA2691 Datasheet, PDF (15/30 Pages) Texas Instruments – Dual Wideband, Current-Feedback OPERATIONAL AMPLIFIER

MACROMODELS AND APPLICATIONS SUPPORT
Computer simulation of circuit performance using SPICE is
often useful when analyzing the performance of analog
circuits and systems. This is particularly true for Video and
RF amplifier circuits where parasitic capacitance and induc-
tance can have a major effect on circuit performance. A
SPICE model for the OPA2691 is available through the TI
web site (www.ti.com). These models do a good job of
predicting small-signal AC and transient performance under
a wide variety of operating conditions. They do not do as well
in predicting the harmonic distortion or dG/dP characteristics.
These models do not attempt to distinguish between the
package types in their small-signal AC performance, nor do
they attempt to simulate channel-to-channel coupling.
OPERATING SUGGESTIONS
SETTING RESISTOR VALUES TO
OPTIMIZE BANDWIDTH
A current-feedback op amp like the OPA2691 can hold an
almost constant bandwidth over signal gain settings with the
proper adjustment of the external resistor values. This is
shown in the Typical Characteristics; the small-signal band-
width decreases only slightly with increasing gain. Those
curves also show that the feedback resistor has been changed
for each gain setting. The resistor values on the inverting side
of the circuit for a current-feedback op amp can be treated as
frequency response compensation elements while their ra-
tios set the signal gain. Figure 7 shows the small-signal
frequency response analysis circuit for the OPA2691.
V_I
I_ERR
α
R_I
R_G
V_O
Z_(S) I_ERR
R_F
FIGURE 7. Current-Feedback Transfer Function Analysis Circuit.
The key elements of this current-feedback op amp model are:
α → Buffer gain from the noninverting input to the inverting input
R_I → Buffer output impedance
i_ERR → Feedback error current signal
Z(s) → Frequency dependent open-loop transimpedance
gain from i_ERR to V_O
The buffer gain is typically very close to 1.00 and is normally
neglected from signal gain considerations. It will, however, set
the CMRR for a single op amp differential amplifier configura-
tion. For a buffer gain α < 1.0, the CMRR = –20 • log (1 – α)dB.
R_I, the buffer output impedance, is a critical portion of the
bandwidth control equation. The OPA2691 is typically about 37Ω.
A current-feedback op amp senses an error current in the
inverting node (as opposed to a differential input error volt-
age for a voltage-feedback op amp) and passes this on to the
output through an internal frequency-dependent transimped-
ance gain. The Typical Characteristics show this open-loop
transimpedance response. This is analogous to the open-
loop voltage gain curve for a voltage-feedback op amp.
Developing the transfer function for the circuit of Figure 7
gives Equation 1:
V_O
V_I
=
α

^{1 +}
R_F
R_G


1+
R_F
+

R_{I }1+
R_F
R_G


=
1+
α NG
R_F + R_I
NG
Z_(S)
Z_(S)
(1)

NG

≡

^{1 +}
R_F
R_G





This is written in a loop-gain analysis format where the errors
arising from a non-infinite open-loop gain are shown in the
denominator. If Z_(S) were infinite over all frequencies, the
denominator of Equation 1 would reduce to 1 and the ideal
desired signal gain shown in the numerator would be achieved.
The fraction in the denominator of Equation 1 determines the
frequency response. Equation 2 shows this as the loop-gain
equation:
Z_{(S) = Loop Gain}
R_F + R_I NG
(2)
If 20 • log(R_F + NG • R_I) were drawn on top of the open-loop
transimpedance plot, the difference between the two would
be the loop gain at a given frequency. Eventually, Z_(S) rolls off
to equal the denominator of Equation 2 at which point the
loop gain has reduced to 1 (and the curves have intersected).
This point of equality is where the amplifier’s closed-loop
frequency response given by Equation 1 will start to roll off,
and is exactly analogous to the frequency at which the noise
gain equals the open-loop voltage gain for a voltage-feed-
back op amp. The difference here is that the total impedance
in the denominator of Equation 2 may be controlled some-
what separately from the desired signal gain (or NG).
The OPA2691 is internally compensated to give a maximally
flat frequency response for R_F = 402Ω at NG = 2 on ±5V
supplies. Evaluating the denominator of Equation 2 (which is
the feedback transimpedance) gives an optimal target of 476Ω.
As the signal gain changes, the contribution of the NG • R_I term
in the feedback transimpedance will change, but the total can
be held constant by adjusting R_F. Equation 3 gives an approxi-
mate equation for optimum R_F over signal gain:
R_F = 476Ω − NG R_I
(3)
As the desired signal gain increases, this equation will even-
tually predict a negative R_F. A somewhat subjective limit to this
adjustment can also be set by holding R_G to a minimum value
of 20Ω. Lower values will load both the buffer stage at the input
and the output stage if R_F gets too low—actually decreasing
OPA2691
15
SBOS224D
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