MAX1549ETL Datasheet, PDF (26/35 Pages) Maxim Integrated Products – Dual, Interleaved, Fixed-Frequency Step-Down Controller with a Dynamically Adjustable Output

Dual, Interleaved, Fixed-Frequency Step-Down
Controller with a Dynamically Adjustable Output
Maximum Load Current: There are two values to con-
sider. The peak inductor current (I_PEAK) determines the
instantaneous component stresses and filtering require-
ments and thus drives output capacitor selection,
inductor saturation rating, and the design of the cur-
rent-limit circuit. The maximum continuous load current
(I_LOAD(MAX)) determines the thermal stresses and thus
drives the selection of input capacitors, MOSFETs, and
other critical heat-contributing components.
Switching Frequency: This choice determines the
basic tradeoff between size, efficiency, and maximum
input voltage range. The optimum frequency is largely
a function of maximum input voltage, due to MOSFET
switching losses that are proportional to frequency and
V_IN². The optimum frequency is also a moving target,
due to rapid improvements in MOSFET technology that
are making higher frequencies more practical.
Inductor Operating Point: This choice provides trade-
offs between size vs. efficiency and transient response
vs. output ripple. Low inductor values provide better tran-
sient response and smaller size, but also result in lower
efficiency and higher output ripple due to increased rip-
ple currents. The minimum practical inductor value is one
that causes the circuit to operate at the edge of critical
conduction (where the inductor current just touches zero
with every cycle at maximum load). Inductor values lower
than this grant no further size-reduction benefit. The opti-
mum operating point is usually found between 20% and
50% ripple current. When pulse skipping (SKIP low and
light loads), the inductor value also determines the load-
current value at which PFM/PWM switchover occurs.
Inductor Selection
The switching frequency and inductor operating point
determine the inductor value as follows:
L=
( ) ^V_OUT ^V_IN ^{- V}_OUT
V_IN x f_{OSC x} I_LOAD(MAX) x LIR
For example: I_LOAD(MAX) = 5A, V_IN = 12V, V_OUT =
2.5V, f_OSC = 300kHz, 30% ripple current or LIR = 0.3:
L=
2.5V × (12V - 2.5V)
= 4.40µH
12V × 300kHz × 5A × 0.3
Find a low-loss inductor with the lowest possible DC
resistance that fits in the allotted dimensions. Ferrite
cores are often the best choice, although powdered
iron is inexpensive and can work well at 200kHz. The
core must be large enough not to saturate at the peak
inductor current (I_PEAK):
I_PEAK
=
I_LOAD(MAX) ^{⎛⎝⎜}1 +
LIR ⎞
2 ⎠⎟
Most inductor manufacturers provide inductors in stan-
dard values, such as 1.0µH, 1.5µH, 2.2µH, 3.3µH, etc.
Also look for nonstandard values, which can provide a
better compromise in LIR across the input voltage range.
If using a swinging inductor (where the no-load induc-
tance decreases linearly with increasing current), evalu-
ate the LIR with properly scaled inductance values.
Transient Response
The inductor ripple current also impacts transient-
response performance, especially at low V_IN - V_OUT dif-
ferentials. Low inductor values allow the inductor
current to slew faster, replenishing charge removed
from the output-filter capacitors by a sudden load step.
The total output voltage sag is the sum of the voltage
sag while the inductor is ramping up and the voltage
sag before the next pulse can occur:
2
(^{( )} ) ^VSAG
=
L ∆I_LOAD(MAX)
2C_OUT V_IN × D_MAX - V_OUT
( ) + ∆ILOAD(MAX) T - ∆T
C_OUT
where D_MAX is the maximum duty factor (see the
Electrical Characteristics table), T is the cycle period (1 /
f_OSC), ∆T equals V_OUT / V_IN x T when in PWM mode, or L
x 0.2 x I_MAX / (V_IN - V_OUT) when in skip mode. The
amount of overshoot during a full-load to no-load transient
due to stored inductor energy can be calculated as:
( ) _{VSOAR ≈}
2
∆I_LOAD(MAX) L
2C_OUTV_OUT
26 ______________________________________________________________________________________