MAX15002 Datasheet, PDF (20/30 Pages) Maxim Integrated Products – Dual-Output Buck Controller with Tracking/Sequencing

Dual-Output Buck Controller with
Tracking/Sequencing
The Type II compensator’s mid-frequency gain
(approximately 4dB shown here) is designed to com-
pensate for the power modulator’s attenuation at the
desired crossover frequency, f_CO (G_E/A + G_MOD = 0dB
at f_CO). In this example, the power modulator’s inherent
-20dB/decade roll-off above the ESR zero (f_ESR) is
leveraged to extend the active regulation gain-band-
width of the voltage regulator. As shown in Figure 5b,
the net result is a 2x increase in the regulator’s gain
bandwidth while providing greater than 55° of phase
margin (the difference between G_E/A and G_MOD
respective phases at crossover, f_CO).
Other filter schemes pose their own problems. For
instance, when choosing high-quality filter capacitor(s),
e.g., MLCCs, and inductor, with minimal parasitics, the
inherent ESR zero can occur at a much higher frequen-
cy, as shown in Figure 5c.
As with the previous example, the actual gain and
phase response is overlaid on the power modulator’s
asymptotic gain response. One readily observes the
more dramatic gain and phase transition at or near the
power modulator’s resonant frequency, f_LC, versus the
gentler response of the previous example. This is due to
the component’s lower parasitics (OCR and ESR) and
corresponding higher frequency of the inherent ESR
zero frequency. In this example, the desired crossover
frequency occurs below the ESR zero frequency.
In this example, a compensator with an inherent mid-
frequency double-zero response is required to mitigate
the effects of the filter’s double-pole. Such is available
with the Type III topology.
As demonstrated in Figure 5d, the Type III’s mid-fre-
quency double-zero gain (exhibiting a +20dB/dec
slope, noting the compensator’s pole at the origin) is
designed to compensate for the power modulator’s
double-pole -40dB/decade attenuation at the desired
crossover frequency, f_CO (again, G_E/A + G_MOD = 0dB
at f_CO). See Figure 5d.
In the above example, the power modulator’s inherent
(mid-frequency) -40dB/decade roll-off is mitigated by
the mid-frequency double zero’s +20dB/decade gain to
extend the active regulation gain-bandwidth of the volt-
age regulator. As shown in Figure 5d, the net result is
an approximate doubling in the regulator’s gain band-
width while providing greater than 60° of phase margin
(the difference between G_E/A and G_MOD respective
phases at crossover, f_CO).
Design procedures for both Type II and Type III com-
pensators are shown below.
POWER MODULATOR GAIN AND PHASE
RESPONSE WITH LOW-PARASITIC
OUTPUT CAPACITORS (MLCCs)
40
90 ^{MAX15002 fig05c}
20
|G_MOD|
45
0
f_LC
0
-20
< G_MOD
-40
f_ESR
-45
-90
-60
-135
|G_MOD|
ASYMPTOTE
-80
-180
10 100 1k 10k 100k 1M 10M
FREQUENCY (Hz)
Figure 5c. Power Modulator Gain and Phase Response (High-
Quality COUT)
POWER MODULATOR AND TYPE III COMPENSATOR
GAIN AND PHASE RESPONSE WITH LOW
PARASITIC OUTPUT CAPACITORS (MLCCs)
80
270 ^{MAX15002 fig05d}
60
< G_EA
203
40
|G_EA|
f_LC
135
20
68
0
f_CO
0
-20
< G_MOD
-40
|G_MOD| _-68
-135
-60
-203
f_ESR
-80
-270
10 100 1k 10k 100k 1M 10M
FREQUENCY (Hz)
Figure 5d. Power Modulator (High-Quality COUT) and Type III
Compensator Responses
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