OPA2677IDDA Datasheet, PDF (19/38 Pages) Texas Instruments – Dual, Wideband, High Output Current Operational Amplifier

frequency response given by Equation 14 starts to roll off, and
is exactly analogous to the frequency at which the noise gain
equals the open-loop voltage gain for a voltage-feedback op
amp. The difference here is that the total impedance in the
denominator of Equation 15 may be controlled somewhat
separately from the desired signal gain (or NG). The OPA2677
is internally compensated to give a maximally flat frequency
response for R_F = 402Ω at NG = 4 on ±6V supplies. Evaluat-
ing the denominator of Equation 15 (which is the feedback
transimpedance) gives an optimal target of 490Ω. As the
signal gain changes, the contribution of the NG • R_I term in
the feedback transimpedance changes, but the total can be
held constant by adjusting R_F. Equation 16 gives an approxi-
mate equation for optimum R_F over signal gain:
R_F = 490 – NG • R_I
(16)
As the desired signal gain increases, this equation eventually
predicts a negative R_F. A somewhat subjective limit to this
adjustment can also be set by holding R_G to a minimum value
of 20Ω. Lower values load both the buffer stage at the input
and the output stage if R_F gets too low—actually decreasing
the bandwidth. Figure 12 shows the recommended R_F versus
NG for both ±6V and a single +5V operation. The values for
R_F versus gain shown here are approximately equal to the
values used to generate the Typical Characteristics. They
differ in that the optimized values used in the Typical Char-
acteristics are also correcting for board parasitic not consid-
ered in the simplified analysis leading to Equation 16. The
values shown in Figure 12 give a good starting point for
designs where bandwidth optimization is desired.
The total impedance going into the inverting input may be
600
is essential for power-supply ripple rejection, noninverting
input noise current shunting, and to minimize the high-
frequency value for R_I in Figure 11.
INVERTING AMPLIFIER OPERATION
The OPA2677 is a general-purpose, wideband current-feed-
back op amp; most of the familiar op amp application circuits
should be available to the designer. Those dual op amp
applications that require considerable flexibility in the feed-
back element (for example, integrators, transimpedance, and
some filters) should consider a unity-gain stable, voltage-
feedback amplifier such as the OPA2822, because the feed-
back resistor is the compensation element for a current-
feedback op amp. Wideband inverting operation (and espe-
cially summing) is particularly suited to the OPA2677. Figure
13 shows a typical inverting configuration where the I/O
impedances and signal gain from Figure 1 are retained in an
inverting circuit configuration.
In the inverting configuration, two key design considerations
50Ω
Source
V_I
R_G
97.6Ω
R_M
102Ω
+6V
Power-supply
decoupling not
shown.
1/2
V_O
OPA2677
50Ω Load
50Ω
R_F
392Ω
V_{O = –} R_{F = –4}
V_I
R_G
–6V
FIGURE 13. Inverting Gain of –4 with Impedance Matching.
500
+5V
400
300
±5V
200
0
5
10
15
20
25
Noise Gain
FIGURE 12. Feedback Resistor vs. Noise Gain.
used to adjust the closed-loop signal bandwidth. Inserting a
series resistor between the inverting input and the summing
junction increases the feedback impedance (denominator of
Equation 15), decreasing the bandwidth. The internal buffer
output impedance for the OPA2677 is slightly influenced by
the source impedance looking out of the noninverting input
terminal. High-source resistors have the effect of increasing
R_I, decreasing the bandwidth. For those single-supply appli-
cations which develop a midpoint bias at the noninverting
input through high-valued resistors, the decoupling capacitor
must be noted. The first is that the gain resistor (R_G)
becomes part of the signal source input impedance. If input
impedance matching is desired (which is beneficial when-
ever the signal is coupled through a cable, twisted pair, long
PC board trace or other transmission line conductor), it is
normally necessary to add an additional matching resistor to
ground. R_G, by itself, is not normally set to the required input
impedance since its value, along with the desired gain, will
determine an R_F, which may be non-optimal from a fre-
quency response standpoint. The total input impedance for
the source becomes the parallel combination of R_G and R_M.
The second major consideration, touched on in the previous
paragraph, is that the signal source impedance becomes
part of the noise gain equation and has a slight effect on the
bandwidth through Equation 15. The values shown in Figure
12 have accounted for this by slightly decreasing R_F (from
the optimum values) to re-optimize the bandwidth for the
noise gain of Figure 12 (NG = 3.98). In the example of Figure
13, the R_M value combines in parallel with the external 50Ω
source impedance, yielding an effective driving impedance of
50Ω || 102Ω = 33.5Ω. This impedance is added in series with
R_G for calculating the noise gain—which gives NG = 3.98.
This value, along with the inverting input impedance of 22Ω,
are inserted into Equation 16 to get a feedback
OPA2677
19
SBOS126I
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