OPA2846_14 Datasheet, PDF (14/30 Pages) Texas Instruments – Dual, Wideband, Low-Noise, Voltage-Feedback Operational Amplifier

OPA2846
SBOS274C −JUNE 2003 − REVISED AUGUST 2008
This will give a −3dB bandwidth approximately equal to:
Ǹ _{f*3dB +} ǒ_{GBPń2pRFCD}Ǔ_Hz
(2)
The example of Figure 4 will give approximately 44MHz
flat bandwidth using the 0.3pF feedback compensation.
If the total output noise is bandlimited to a frequency less
than the feedback pole frequency, a very simple
expression for the equivalent input noise current can be
derived as:
Ǹ ǒ Ǔ _{IEQ +}
I
2
N
)
4kT
R_F
)
E_N
R_F
2
)
(E_N2pC_DF)²
3
(3)
Where:
I_EQ = Equivalent input noise current if the output noise is
bandlimited to F < 1/(2πR_FC_F)
I_N = Input current noise for the op amp inverting input
E_N = Input voltage noise for the op amp
C_D = Diode capacitance
F = Bandlimiting frequency in Hz (usually a post filter
prior to further signal processing)
Evaluating this expression up to the feedback pole
frequency at 31MHz for the circuit of Figure 4 gives an
equivalent input noise current of 3.1pA/Hz. This is only
slightly higher than the current noise of the op amp itself.
TWO-STAGE TRANSIMPEDANCE DESIGN
The dual OPA2846 may be used as either a dual
transimpedance channel from two photodetectors, or as a
very high gain stage by using one amplifier as the
transimpedance stage with the second used as a post gain
amplifier. See Figure 5 for an example of using one
channel as a transimpedance front end from a large area
detector, with the second amplifier used as a voltage gain
stage to get a 100kΩ total gain (Z_T) from a large 50pF
detector (C_D in Figure 5).
1000pF
2.67kΩ
1/2
O PA 2 84 6
λ
C_D
50pF
− V_B
2.67kΩ
1.9pF
20Ω
1 /2
O P A 28 46
732Ω
20Ω
Figure 5. High-Gain, Wideband Transimpedance
Amplifier
14
www.ti.com
One key question in this design is how best to split up the
first and second stage gains. If bandwidth optimization
from a given photodetector capacitance (C_D in Figure 5) is
the primary goal, Equation 4 gives a solution for R_F in the
input stage that will provide an equal bandwidth in the first
and second stages, giving the maximum overall channel
bandwidth.
2
ȧȡȢ ȧȣȤ ^RF ⁺
Z_T
2pC_D GBP
(4)
Where:
Z_T = Desired total transimpedance gain
C_D = Diode capacitance at reverse bias
GBP = Amplifier Gain Bandwidth Product (MHz)
This equation is used to calculate the required input stage
feedback resistor in Figure 5. The remaining total signal
gain is provided by the second stage; in the example of
Figure 5, setting G = 37.5 gives the same bandwidth
(approximately 44MHz) as the bandwidth achieved by the
input stage. To set this first stage bandwidth to its
maximally flat values, use Equation 5 to set the feedback
capacitor value:
Ǹǒ Ǔ C_F +
C_D
p R_F GBP
(5)
f _*3dB
+
1
Ǹ₂
(GBP)^2ń3
(2pC_D)^1ń3 (Z_T)^1ń3
(6)
The approximate achievable bandwidth in the two stages
is given by Equation 6, which gives approximately 30MHz
for Figure 5.
LOW-GAIN COMPENSATION FOR
IMPROVED SFDR
Where a low gain is desired, and inverting operation is
acceptable, a new external compensation technique may
be used to retain the full slew rate and noise benefits of the
OPA2846 while giving increased loop gain and the
associated improvement in distortion offered by the
decompensated architecture. This technique shapes the
loop gain for good stability while giving an easily-con-
trolled, 2nd-order, low-pass frequency response. Consid-
ering only the noise gain (noninverting signal gain, which
is also called the Noise Gain or NG) for the circuit of
Figure 6, the low-frequency noise gain, (NG₁) will be set
by the resistor ratios while the high-frequency noise gain
(NG₂) will be set by the capacitor ratios. The capacitor
values set both the transition frequencies and the
high-frequency noise gain. If this noise gain (determined
by NG₂ = 1 + C_S/C_F) is set to a value greater than the
recommended minimum stable gain for the op amp, and
the noise gain pole (set by 1/R_FC_F) is placed correctly, a
very well-controlled, 2nd-order, low-pass frequency
response will result.