MAX15035 Datasheet, PDF (20/27 Pages) Maxim Integrated Products – 15A Step-Down Regulator with Internal Switches

15A Step-Down Regulator with Internal Switches
• Switching frequency: This choice determines the
basic trade-off between size and efficiency. The
optimal frequency is largely a function of maximum
input voltage due to MOSFET switching losses that
are proportional to frequency and V_IN². The opti-
mum frequency is also a moving target, due to
rapid improvements in MOSFET technology that are
making higher frequencies more practical.
• Inductor operating point: This choice provides
trade-offs between size vs. efficiency and transient
response vs. output noise. Low inductor values pro-
vide better transient response and smaller physical
size, but also result in lower efficiency and higher
output noise due to increased ripple current. The
minimum practical inductor value is one that causes
the circuit to operate at the edge of critical conduc-
tion (where the inductor current just touches zero
with every cycle at maximum load). Inductor values
lower than this grant no further size-reduction bene-
fit. The optimum operating point is usually found
between 20% and 50% ripple current.
Inductor Selection
The switching frequency and operating point (% ripple
current or LIR) determine the inductor value as follows:
L
=
⎛
⎝⎜
V_IN − V_OUT
f_SWI_LOAD(MAX)LIR
⎞
⎠⎟
⎛
⎝⎜
V_OUT
V_IN
⎞
⎠⎟
Find a low-loss inductor having the lowest possible DC
resistance that fits in the allotted dimensions. Ferrite
cores are often the best choice, although powdered
iron is inexpensive and can work well at 200kHz. The
core must be large enough not to saturate at the peak
inductor current (I_PEAK):
I_PEAK
=
I_LOAD(MAX)
+
∆I_L
2
Transient Response
The inductor ripple current impacts transient-response
performance, especially at low V_IN - V_OUT differentials.
Low inductor values allow the inductor current to slew
faster, replenishing charge removed from the output fil-
ter capacitors by a sudden load step. The amount of
output sag is also a function of the maximum duty factor,
which can be calculated from the on-time and minimum
off-time. The worst-case output sag voltage can be
determined by:
( ) _VSAG
=
L ∆I_LOAD(MAX) ²
2C_OUTV_OUT
⎡⎛
⎣⎢⎢⎝⎜⎜
(^V_IN
⎡⎛
⎢⎜
⎣⎝
V_OUTt_SW
V_IN
⎞
⎟
⎠
+
t_OFF(MIN)
⎤
⎥
⎦
−
V_OUT
V_IN
)
t_SW
⎞
⎠⎟⎟
−
t_OFF(MIN)
⎤
⎥
⎦⎥
where t_OFF(MIN) is the minimum off-time (see the Electrical
Characteristics table).
The amount of overshoot due to stored inductor energy
when the load is removed can be calculated as:
( ) _{VSOAR ≈}
∆I_LOAD(MAX) ²L
2C_OUTV_OUT
Setting the Valley Current Limit
The minimum current-limit threshold must be high
enough to support the maximum load current when the
current limit is at the minimum tolerance value. The val-
ley of the inductor current occurs at I_LOAD(MAX) minus
half the inductor ripple current (∆I_L); therefore:
I_LIMIT(LOW)
>
I_LOAD(MAX)
−
∆I_L
2
where I_LIMIT(LOW) equals the minimum current-limit
threshold voltage divided by 0.006.
The valley current-limit threshold is precisely 1/20 the
voltage seen at ILIM. Connect a resistive divider from
REF to ILIM to analog ground (AGND) to set a fixed val-
ley current-limit threshold. The external 400mV to 2V
adjustment range corresponds to a 20mV to 100mV val-
ley current-limit threshold. When adjusting the current-
limit threshold, use 1% tolerance resistors and a divider
current of approximately 5µA to 10µA to prevent signifi-
cant inaccuracy in the valley current-limit tolerance.
The MAX15035 uses the low-side MOSFET’s on-resis-
tance as the current-sense element (R_SENSE =
R_DS(ON)). A good general rule is to allow 0.5% addi-
tional resistance for each degree celsius of tempera-
ture rise, which must be included in the design margin
unless the design includes an NTC thermistor in the
ILIM resistive voltage-divider to thermally compensate
the current-limit threshold.
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