MAX1632AEAI Datasheet, PDF(19/29 Page) Maxim Integrated Products – Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers

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MAX1632AEAI Datasheet, PDF (19/29 Pages) Maxim Integrated Products – Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers

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Multi-Output, Low-Noise Power-Supply

Controllers for Notebook Computers

Current-Sense Resistor Value

The current-sense resistor value is calculated according

to the worst-case-low current-limit threshold voltage

(from the Electrical Characteristics table) and the peak

inductor current:

R SENSE

80mV

IPEAK

Use IPEAK from the second equation in the Inductor

Value section

Use the calculated value of RSENSE to size the MOSFET

switches and specify inductor saturation-current ratings

according to the worst-case high-current-limit threshold

voltage:

IPEAK(MAX)

120mV

RSENSE

Output Filter Capacitor Value

The output filter capacitor values are generally deter-

mined by the ESR and voltage rating requirements, rather

than actual capacitance requirements for loop stability. In

other words, the low-ESR electrolytic capacitor that meets

the ESR requirement usually has more output capaci-

tance than is required for AC stability. Use only special-

ized low-ESR capacitors intended for switching-regulator

applications, such as AVX TPS, Sprague 595D, Sanyo

OS-CON, or Nichicon PL series. To ensure stability, the

capacitor must meet both minimum capacitance and

maximum ESR values as given in the following equations:

COUT >

VREF(1 + VOUT / VIN(MIN))

VOUT x RSENSE x f

RESR <

RSENSE x VOUT

VREF

Low-inductance resistors, such as surface-mount

metal-film, are recommended.

Input Capacitor Value

Connect low-ESR bulk capacitors and small ceramic

capacitors (0.1ÂµF) directly to the drains on the high-side

MOSFETs. The bulk input filter capacitor is usually

selected according to input ripple current requirements

and voltage rating, rather than capacitor value.

Electrolytic capacitors with low enough effective series

resistance (ESR) to meet the ripple current requirement

invariably have sufficient capacitance values. Aluminum

electrolytic capacitors, such as Sanyo

OS-CON or Nichicon PL, are superior to tantalum types,

which carry the risk of power-up surge-current failure,

especially when connecting to robust AC adapters or

low-impedance batteries. RMS input ripple current

(IRMS) is determined by the input voltage and load cur-

rent, with the worst case occurring at VIN = 2 x VOUT:

IRMS = ILOAD x

VOUT(VIN - VOUT)

VIN

Therefore, when VIN is 2 X VOUT:

IRMS =

ILOAD

Bypassing V+

Bypass the V+ input with a 4.7ÂµF tantalum capacitor

paralleled with a 0.1ÂµF ceramic capacitor, close to the

IC. A 10Î© series resistor to VIN is also recommended.

Bypassing VL

Bypass the VL output with a 4.7ÂµF tantalum capacitor

paralleled with a 0.1ÂµF ceramic capacitor, close to the

device.

(can be multiplied by 1.5; see text below)

These equations are worst case, with 45 degrees of

phase margin to ensure jitter-free, fixed-frequency

operation and provide a nicely damped output

response for zero to full-load step changes. Some cost-

conscious designers may wish to bend these rules with

less-expensive capacitors, particularly if the load lacks

large step changes. This practice is tolerable if some

bench testing over temperature is done to verify

acceptable noise and transient response.

No well-defined boundary exists between stable and

unstable operation. As phase margin is reduced, the

first symptom is a bit of timing jitter, which shows up as

blurred edges in the switching waveforms where the

scope does not quite sync up. Technically speaking,

this jitter (usually harmless) is unstable operation, since

the duty factor varies slightly. As capacitors with higher

ESRs are used, the jitter becomes more pronounced,

and the load-transient output voltage waveform starts

looking ragged at the edges. Eventually, the load-tran-

sient waveform has enough ringing on it that the peak

noise levels exceed the allowable output voltage toler-

ance. Note that even with zero phase margin and gross

instability present, the output voltage noise never gets

much worse than IPEAK x RESR (under constant loads).

Designers of RF communicators or other noise-sensi-

tive analog equipment should be conservative and stay

within the guidelines. Designers of notebook computers

and similar commercial-temperature-range digital

systems can multiply the RESR value by a factor of 1.5

without hurting stability or transient response.

The output voltage ripple is usually dominated by the

filter capacitorâs ESR, and can be approximated as

IRIPPLE x RESR. There is also a capacitive term, so the

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