LTC3890-2 Datasheet, PDF (19/40 Pages) Linear Technology – 60V Low IQ, Dual, 2-Phase Synchronous Step-Down DC/DC Controller

LTC3890-2
APPLICATIONS INFORMATION
The inductor value has a direct effect on ripple current. The
inductor ripple current, ΔI_L, decreases with higher induc-
tance or higher frequency and increases with higher V_IN:
ΔI_L
=
1
(^f)(^L)
V_OUT
⎛
⎜^1–
⎝
V_OUT
V_IN
⎞
⎟
⎠
Accepting larger values of ΔI_L allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ΔI_L = 0.3(I_MAX). The maximum
ΔI_L occurs at the maximum input voltage.
The inductor value also has secondary effects. The tran-
sition to Burst Mode operation begins when the average
inductor current required results in a peak current below
25% of the current limit determined by R_SENSE. Lower
inductor values (higher ΔI_L) will cause this to occur at
lower load currents, which can cause a dip in efficiency in
the upper range of low current operation. In Burst Mode
operation, lower inductance values will cause the burst
frequency to decrease.
Inductor Core Selection
Once the value for L is known, the type of inductor must
be selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite or molypermalloy
cores. Actual core loss is independent of core size for a
fixed inductor value, but it is very dependent on inductance
value selected. As inductance increases, core losses go
down. Unfortunately, increased inductance requires more
turns of wire and therefore copper losses will increase.
Ferrite designs have very low core loss and are preferred
for high switching frequencies, so design goals can con-
centrate on copper loss and preventing saturation. Ferrite
core material saturates hard, which means that induc-
tance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Power MOSFET and Schottky Diode
(Optional) Selection
Two external power MOSFETs must be selected for each
controller in the LTC3890-2: one N-channel MOSFET for
the top (main) switch, and one N-channel MOSFET for the
bottom (synchronous) switch.
The peak-to-peak drive levels are set by the INTV_CC voltage.
This voltage is typically 5.1V during start-up (see EXTV_CC
Pin Connection). Consequently, logic-level threshold
MOSFETs must be used in most applications. Pay close
attention to the BV_DSS specification for the MOSFETs as well.
Selection criteria for the power MOSFETs include the
on-resistance, R_DS(ON), Miller capacitance, C_MILLER, input
voltage and maximum output current. Miller capacitance,
C_MILLER, can be approximated from the gate charge curve
usually provided on the MOSFET manufacturers’ data
sheet. C_MILLER is equal to the increase in gate charge
along the horizontal axis while the curve is approximately
flat divided by the specified change in V_DS. This result is
then multiplied by the ratio of the application applied V_DS
to the gate charge curve specified V_DS. When the IC is
operating in continuous mode the duty cycles for the top
and bottom MOSFETs are given by:
Main Switch Duty Cycle = ^VOUT
V_IN
Synchronous Switch Duty Cycle = ^{VIN − VOUT}
V_IN
The MOSFET power dissipations at maximum output
current are given by:
P_MAIN
=
V_OUT
V_IN
(^I_MAX )²
(¹⁺ )^{δ R}_DS(ON)
+
(^V_IN
)²
⎛⎝⎜I_M2AX
⎞
⎠⎟
(^R_DR
)
(^C_MILLER
)
•
⎡
⎢
1
+
1
⎤
⎥(^f)
⎣ ^V_INTVCC ^{– V}_THMIN ^V_THMIN ⎦
P_SYNC
=
V_IN
– V_OUT
V_IN
(^I_MAX )²
(¹⁺
)^{δ R}_DS(ON)
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