OPA2686 Datasheet, PDF (10/18 Pages) Burr-Brown (TI) – Dual, Wideband, Low Noise, Voltage Feedback OPERATIONAL AMPLIFIER

tion improved through the use of a diplexer line interface.
The differential receive signal is brought into the inverting
channel gain resistors to get both noise and distortion
improvement for a given desired gain setting. To get
impedance matching, set 2R_G equal to the required load
looking out of the diplexer. The signal gain is then set by
adjusting feedback resistors, R_F. Using the OPA2686 in the
inverting mode will give you a reduced noise gain as
described in the “Wideband, Inverting Gain Operation”
section of this data sheet. This will improve both the SNR
and distortion performance. If the noise gain for a particular
application drops below the minimum recommended stable
gain (+7), consider using the Low Gain Compensation
technique described later in this data sheet.
SINGLE-STAGE TRANSIMPEDANCE DESIGN
When setting up either one or both stages as a broadband
photodiode amplifier, the key elements in the design are the
expected diode capacitance (C_D) with the reverse bias volt-
age (–V_B) applied, the desired transimpedance gain R_F, and
the GBP of the OPA2686 (1600MHz). Figure 3 shows a
design using a 10pF source capacitance diode and a 10kΩ
transimpedance gain. With these three variables set (and
including the parasitic input capacitance for the OPA2686
added to C_D), the feedback capacitor value (C_F) may be set
to control the frequency response.
+5V
Supply Decoupling
Not Shown
1/2
OPA2686
V_O = I_D R_F
R_F
10kΩ
λ
I_D
C_D
10pF
C_F
–5V _0.3pF
–V_B
FIGURE 3. Wideband, Low Noise, Transimpedance
Amplifier.
a parasitic capacitance of 0.2pF, leaving the required 0.3pF
value shown in Figure 3 to get the required feedback pole.
This will give a –3dB bandwidth approximately equal to:
f_–3dB = √(GBP/2πR_FC_D)Hz
Eq. 2
The example of Figure 3 will give approximately 44MHz
flat bandwidth using the 0.3pF feedback compensation.
If the total output noise is bandlimited to a frequency less
than the feedback pole frequency, a very simple expression
for the equivalent input noise current can be derived as:
( ) _{IEQ =}
I ^2N
+
4kT
R_F
+


E_N
R_F


2
+
E_N 2πC_DF
3
2
Eq. 3
Where:
I_EQ = Equivalent input noise current if the output noise is
bandlimited to F < 1/(2πR_FC_D)
I_N = Input current noise for the op amp inverting input
E_N = Input voltage noise for the op amp
C_D = Diode capacitance
F = Bandlimiting frequency in Hz (usually a post filter
prior to further signal processing)
Evaluating this expression up to the feedback pole fre-
quency at 31MHz for the circuit of Figure 3 gives an
equivalent input noise current of 2.6pA/√Hz. This is only
slightly higher than the current noise of the op amp itself.
TWO-STAGE TRANSIMPEDANCE DESIGN
The dual OPA2686 may be used as either a dual
transimpedance channel from two photodectors or as a very
high gain stage by using one amplifier as the transimpedance
stage with the second used as a post gain amplifier. Figure
4 shows an example of using one channel as a transimpedance
front end from a large area detector, with the second ampli-
fier used as a voltage gain stage to get a 100kΩ total gain
(Z_T) from a large 50pF detector, (C_D in Figure 4).
One key question in this design is how best to split up the
first and second stage gains. If bandwidth optimization from
a given photodetector capacitance (C_D in Figure 4) is the
To achieve a maximally flat 2nd-order Butterworth fre-
quency response, the feedback pole should be set to:
1/(2πR_FC_F) = √(GBP/(4πR_FC_D))
Eq. 1
0.1µF
2.67kΩ
1/2
OPA2686
20Ω
1/2
OPA2686
Adding the common-mode and differential mode input ca-
pacitance (1.0 + 2.0)pF to the 10pF diode source capacitance
λ
of Figure 3, and targeting a 10kΩ transimpedance gain using
the 1600MHz GBP for the OPA2686, will require a feed-
C_D
50pF
2.67kΩ
1.9pF
732Ω
20Ω
back pole set to 31MHz. This will require a total feedback
–V_B
capacitance of 0.5pF. Typical surface-mount resistors have
FIGURE 4. High Gain, Wideband Transimpedance Amplifier.
®
OPA2686
10