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OPA683IDBVR Datasheet, PDF (16/31 Pages) Texas Instruments – Very Low-Power, Current Feedback RATIONAL AMPLIFIER With Disable
◁
A current-feedback op amp senses an error current in the
inverting node (as opposed to a differential input error volt-
age for a voltage feedback op amp) and passes this on to the
output through an internal frequency dependent transimped-
ance gain. The Typical Characteristics show this open-loop
transimpedance response. This is analogous to the open-
loop voltage gain curve for a voltage feedback op amp.
Developing the transfer function for the circuit of Figure 10
gives Equation 1:
(1)
VO
VI
=
α

1+
RF
RG


1+
RF
+
RI
1+
RF
RG


=
1+
α NG
RF + RI
NG
Z(S)
Z(S)

NG

=

1
+
RF
RG





This is written in a loop-gain analysis format where the errors
arising from a non-infinite open-loop gain are shown in the
denominator. If Z(s) was infinite over all frequencies, the
denominator of Equation 1 would reduce to 1 and the ideal
desired signal gain shown in the numerator would be achieved.
The fraction in the denominator of Equation 1 determines the
frequency response. Equation 2 shows this as the loop-gain
equation.
(2)
Z(S)
= Loop Gain
RF + RI NG
If 20 • log(RF + NG • RI) were drawn on top of the open-loop
transimpedance plot, the difference between the two would
be the loop gain at a given frequency. Eventually, Z(s) rolls
off to equal the denominator of Equation 2 at which point the
loop gain has reduced to 1 (and the curves have intersected).
This point of equality is where the amplifier’s closed-loop
frequency response given by Equation 1 will start to roll off,
and is exactly analogous to the frequency at which the noise
gain equals the open-loop voltage gain for a voltage feed-
back op amp. The difference here is that the total impedance
in the denominator of Equation 2 may be controlled some-
what separately from the desired signal gain (or NG).
The OPA683 is internally compensated to give a maximally
flat frequency response for RF = 1.2kΩ at NG = 2 on ±5V
supplies. That optimum value goes to 1.4kΩ on a single +5V
supply. Normally, with a current feedback amplifier, it is
possible to adjust the feedback resistor to hold this band-
width up as the gain is increased. The CFBplus architecture
has reduced the contribution of the inverting input impedance
to provide exceptional bandwidth to higher gains without
adjusting the feedback resistor value. The Typical Character-
istics show the small-signal bandwidth over gain with a fixed
feedback resistor.
At very high gains, 2nd-order effects in the buffer output
impedance cause the overall response to peak up. If desired,
it is possible to retain a flatter frequency response at higher
gains by adjusting the feedback resistor to higher values as
the gain is increased. Figure 11 shows the empirically deter-
mined feedback resistor and resulting –3dB bandwidth from
gains of +2 to +100 to hold a < 0.5dB peaked response.
Here, since a slight peaking was allowed, a lower nominal RF
is suggested at a gain of +2 giving > 250MHz bandwidth.
This exceeds that shown in the Electrical Characteristics due
to the slightly lower feedback resistor allowing a modest
peaking in the response. Figure 12 shows the measured
frequency response curves with the adjusted feedback resis-
tor value. While the bandwidth for this low-power part does
reduce at higher gains, going over a 50:1 gain range gives
only a factor of 10 bandwidth reduction. The 25MHz band-
width at a gain of 100V/V is equivalent to a 2.5GHz gain
bandwidth product voltage feedback amplifier capability. Even
better bandwidth retention to higher gains can be delivered
by the slightly higher quiescent power OPA684.
3900
3400
2900
2400
1900
1400
900
VO = 0.5VPP
–3dB Bandwidth
RF
2
5
10
20
50
Voltage Gain (V/V)
325
275
225
175
125
75
25
100
FIGURE 11. Bandwidth and RF Optimized vs Gain.
3
0
–3
–6
–9
–12
1
G=5
G=2
G = 100
G = 50
G = 20
10
Frequency (MHz)
G = 10
100 200
FIGURE 12. Small-Signal Frequency Response with Opti-
mized RF.
16
OPA683
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