SA5212A Datasheet, PDF (11/20 Pages) NXP Semiconductors – Transimpedance amplifier 140MHz

Philips Semiconductors
Transimpedance amplifier (140MHz)
Product specification
SA5212A
THEORY OF OPERATION
Transimpedance amplifiers have been widely used as the
preamplifier in fiber-optic receivers. The SA5212A is a wide
bandwidth (typically 140MHz) transimpedance amplifier designed
primarily for input currents requiring a large dynamic range, such as
those produced by a laser diode. The maximum input current before
output stage clipping occurs at typically 240µA. The SA5212A is a
bipolar transimpedance amplifier which is current driven at the input
and generates a differential voltage signal at the outputs. The
forward transfer function is therefore a ratio of the differential output
voltage to a given input current with the dimensions of ohms. The
main feature of this amplifier is a wideband, low-noise input stage
which is desensitized to photodiode capacitance variations. When
connected to a photodiode of a few picoFarads, the frequency
response will not be degraded significantly. Except for the input
stage, the entire signal path is differential to provide improved
power-supply rejection and ease of interface to ECL type circuitry. A
block diagram of the circuit is shown in Figure 10. The input stage
(A1) employs shunt-series feedback to stabilize the current gain of
the amplifier. The transresistance of the amplifier from the current
source to the emitter of Q₃ is approximately the value of the
feedback resistor, R_F=7kΩ. The gain from the second stage (A2)
and emitter followers (A3 and A4) is about two. Therefore, the
differential transresistance of the entire amplifier, R_T is
R_T
+
V_OUT(diff)
I_IN
+ 2R_F
+ 2(7.2K)
+ 14.4kW
The single-ended transresistance of the amplifier is typically 7.2kΩ.
The simplified schematic in Figure 11 shows how an input current is
converted to a differential output voltage. The amplifier has a single
input for current which is referenced to Ground 1. An input current
from a laser diode, for example, will be converted into a voltage by
the feedback resistor R_F. The transistor Q1 provides most of the
open loop gain of the circuit, A_VOL≈70. The emitter follower Q₂
minimizes loading on Q₁. The transistor Q₄, resistor R₇, and V_B1
provide level shifting and interface with the Q₁₅ – Q₁₆ differential
pair of the second stage which is biased with an internal reference,
V_B2. The differential outputs are derived from emitter followers Q₁₁ –
Q₁₂ which are biased by constant current sources. The collectors of
Q₁₁ – Q₁₂ are bonded to an external pin, V_CC2, in order to reduce
the feedback to the input stage. The output impedance is about 17Ω
single-ended. For ease of performance evaluation, a 33Ω resistor is
used in series with each output to match to a 50Ω test system.
BANDWIDTH CALCULATIONS
The input stage, shown in Figure 12, employs shunt-series feedback
to stabilize the current gain of the amplifier. A simplified analysis can
determine the performance of the amplifier. The equivalent input
capacitance, C_IN, in parallel with the source, I_S, is approximately
7.5pF, assuming that C_S=0 where C_S is the external source
capacitance.
Since the input is driven by a current source the input must have a
low input resistance. The input resistance, R_IN, is the ratio of the
incremental input voltage, V_IN, to the corresponding input current, I_IN
and can be calculated as:
R_IN
+
V_IN
I_IN
+
1
R_F
) A_VOL
+
7.2K
70
+ 103W
More exact calculations would yield a higher value of 110Ω.
Thus C_IN and R_IN will form the dominant pole of the entire amplifier;
f_*3dB
+
2p
1
R_IN
C_IN
Assuming typical values for R_F = 7.2kΩ, R_IN = 110Ω, C_IN = 10pF
f_*3dB
+
2p
(110)
1
10 @ 10^*12
+ 145MHz
The operating point of Q1, Figure 2, has been optimized for the
lowest current noise without introducing a second dominant pole in
the pass-band. All poles associated with subsequent stages have
been kept at sufficiently high enough frequencies to yield an overall
single pole response. Although wider bandwidths have been
achieved by using a cascade input stage configuration, the present
solution has the advantage of a very uniform, highly desensitized
frequency response because the Miller effect dominates over the
external photodiode and stray capacitances. For example, assuming
a source capacitance of 1pF, input stage voltage gain of 70, R_IN =
60Ω then the total input capacitance, C_IN = (1+7.5) pF which will
lead to only a 12% bandwidth reduction.
OUTPUT +
A3
INPUT
A1
A2
R_F
A4
OUTPUT –
SD00327
Figure 10. SA5212A – Block Diagram
NOISE
Most of the currently installed fiber-optic systems use non-coherent
transmission and detect incident optical power. Therefore, receiver
noise performance becomes very important. The input stage
achieves a low input referred noise current (spectral density) of
3.5pA/√Hz. The transresistance configuration assures that the
external high value bias resistors often required for photodiode
biasing will not contribute to the total noise system noise. The
equivalent input _RMS noise current is strongly determined by the
quiescent current of Q₁, the feedback resistor R_F, and the
bandwidth; however, it is not dependent upon the internal
Miller-capacitance. The measured wideband noise was 52nA RMS
in a 200MHz bandwidth.
1998 Oct 07
11