LTC3809-1_15 Datasheet, PDF (12/24 Pages) Linear Technology – No RSENSE, Low Input Voltage, Synchronous DC/DC Controller with Output Tracking

LTC3809-1
APPLICATIONS INFORMATION
The typical LTC3809-1 application circuit is shown in Figure
8. External component selection for the controller is driven
by the load requirement and begins with the selection of
the inductor and the power MOSFETs.
Power MOSFET Selection
The LTC3809-1’s controller requires two external power
MOSFETs: a P-channel MOSFET for the topside (main)
switch and a N-channel MOSFET for the bottom (synchro-
nous) switch. The main selection criteria for the power
MOSFETs are the breakdown voltage V_BR(DSS), threshold
voltage V_GS(TH), on-resistance R_DS(ON), reverse transfer
capacitance C_RSS, turn-off delay t_D(OFF) and the total gate
charge Q_G.
The gate drive voltage is the input supply voltage. Since
the LTC3809-1 is designed for operation down to low input
voltages, a sublogic level MOSFET (R_DS(ON) guaranteed at
V_GS = 2.5V) is required for applications that work close to
this voltage. When these MOSFETs are used, make sure that
the input supply to the LTC3809-1 is less than the absolute
maximum MOSFET V_GS rating, which is typically 8V.
The P-channel MOSFET’s on-resistance is chosen based
on the required load current. The maximum average load
current I_OUT(MAX) is equal to the peak inductor current
minus half the peak-to-peak ripple current I_RIPPLE. The
LTC3809-1’s current comparator monitors the drain-to-
source voltage V_DS of the top P-channel MOSFET, which
is sensed between the V_IN and SW pins. The peak inductor
current is limited by the current threshold, set by the voltage
on the I_TH pin, of the current comparator. The voltage on
the I_TH pin is internally clamped, which limits the maximum
current sense threshold ΔV_SENSE(MAX) to approximately
125mV when IPRG is floating (85mV when IPRG is tied
low; 204mV when IPRG is tied high).
The output current that the LTC3809-1 can provide is
given by:
I_OUT(MAX)
=
ΔV_SENSE(MAX)
R_DS(ON)
–
I_RIPPLE
2
where I_RIPPLE is the inductor peak-to-peak ripple current
(see Inductor Value Calculation).
A reasonable starting point is setting ripple current I_RIPPLE
to be 40% of I_OUT(MAX). Rearranging the above equation
yields:
R_DS(ON)MAX
=
5
6
•
ΔV_SENSE(MAX)
I_OUT(MAX)
for
Duty
Cycle <
20%
However, for operation above 20% duty cycle, slope
compensation has to be taken into consideration to select
the appropriate value of R_DS(ON) to provide the required
amount of load current:
R_DS(ON)MAX
=
5
6
•
SF
•
ΔV_SENSE(MAX)
I_OUT(MAX)
where SF is a scale factor whose value is obtained from
the curve in Figure 1.
These must be further derated to take into account the
significant variation in on-resistance with temperature. The
following equation is a good guide for determining the re-
quired R_DS(ON)MAX at 25°C (manufacturer’s specification),
allowing some margin for variations in the LTC3809-1 and
external component values:
R_DS(ON)MAX
=
5
6
•
0.9
•
SF
•
ΔV_SENSE(MAX)
I_OUT(MAX) • ρ_T
The ρ_T is a normalizing term accounting for the temperature
variation in on-resistance, which is typically about 0.4%/°C,
as shown in Figure 2. Junction-to-case temperature T_JC is
about 10°C in most applications. For a maximum ambi-
ent temperature of 70°C, using ρ_80°C ~ 1.3 in the above
equation is a reasonable choice.
The N-channel MOSFET’s on resistance is chosen based
on the short-circuit current limit (I_SC). The LTC3809-
1’s short-circuit current limit comparator monitors the
drain-to-source voltage V_DS of the bottom N-channel
MOSFET, which is sensed between the GND and SW pins.
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