HMC661LC4B Datasheet, PDF (9/14 Pages) Hittite Microwave Corporation – ULTRA-WIDEBAND 4 GS/s TRACK-AND-HOLD AMPLIFIER

HMC661LC4B
v00.0312
ULTRA-WIDEBAND 4 GS/s TRACK-AND-HOLD AMPLIFIER
DC - 18 GHz
Linearity Measurement (Continued)
the track-mode portion of the T/H 1 output and permits a spectrum analyzer linearity measurement of the cascaded
devices. Since T/H 2 only has to sample the held waveform from T/H 1, the linearity impact of T/H 2 is primarily
associated with its DC linearity. An often used approximation is that the DC linearity of T/H 2 is much higher than the
slew-rate dependent, high frequency linearity of T/H 1 so that the total non-linearity of the cascade is dominated by
the high frequency linearity of T/H 1. In this case, the dual rank configuration has a net linearity that closely resembles
the linearity of a single T/H, particularly at high frequencies. However, this approximation is not always valid. If not, the
dual rank configuration fails to represent the linearity of a single T/H. The HMC661LC4B represents such a case; the
3rd order nonlinearity of this device varies relatively slowly with frequency and is high enough over the T/H bandwidth
that the DC linearity of the 2nd T/H significantly impacts the overall dual rank configuration for signals frequencies <5
GHz.
In order to address the weakness of the dual rank technique for the single rank T/H linearity characterization, Hittite
has developed a modified approach referred to as the attenuated dual rank technique. This method involves using
a dual rank T/H configuration with significant attenuation A(dB) inserted between the 1st T/H and the 2nd T/H. A
typical attenuation value is A = 10 dB although more can be used for improved linearity measurement accuracy at
the expense of measurement dynamic range. This configuration still outputs a dual rank waveform that eliminates
the track mode component but the attenuation of the signal into the 2nd device substantially reduces the contribution
of its nonlinear products to the total spectrum such that the linearity of the 1st device dominates the overall linearity.
This results from the fact that the device follows the normal nonlinear order dependence: the 2nd order product level
relative to the fundamental is lowered by 10 dB for every 10 dB decrease in input level while the 3rd order product
level relative to the fundamental is lowered by 20 dB for every 10 dB decrease in input level. The phase of the T/H 2
nonlinear products relative to the T/H 1 products can vary depending on the product frequency and the path length
between the two devices. A worst case analysis of the uncertainty generated by the presence of the lower level T/H
2 products assuming identical devices for the two T/Hs and worst case phasing can be performed. Under these
conditions, the worst case error bounds in the determination of T/H 1 spurious-free dynamic range (SFDR,) due to the
presence of lower level T/H 2 non linear products is given by:
∆SFDR2(dB) = 20 log (1±10 ^-A/20) = +2.4, -3.3 dB for A = 10 dB
∆SFDR3(dB) = 20 log (1±10 ^-A/10) = +0.83, -0.92 dB for A = 10 dB
Where ∆SFDR2 and ∆SFDR3 are the errors in the 2nd order and 3rd order limited spurious-free dynamic ranges for
T/H 1 respectively.
Over the range of DC - 5 GHz the 3rd order products tend to dominate the spurious-free dynamic range so the typical
accuracy in assessing T/H 1 SFDR is of the order of ±0.9 dB for this measurement method. At higher frequencies, the
total linearity is dominated by the high frequency nonlinearity of T/H 1 and the contribution of T/H 2 DC nonlinearity is
much lower than that indicated by ∆SFDR2 above (which simplistically assumes equal linearity across the bandwidth).
So the T/H 1 linearity determination error over the entire frequency range is of the order ±0.9 dB for the attenuated
dual rank measurement method with A = 10 dB.
Another linearity measurement issue unique to the T/H device is the need for output waveform frequency response
correction. In the case of a dual rank T/H, the output waveform resembles a square wave with duration equal to the
clock period. Mathematically, the output can be viewed as the convolution of an ideal delta-function sample train
with a single square pulse of duration equal to one clock period. This weights the output spectral content with a
SIN(πf/fs )/(πf/fs) (Sinc) function frequency response envelope which has nulls at harmonics of the clock frequency fs
and substantial response reduction beyond half the clock frequency. This spectral content and envelope function are
observed during spectrum analyzer measurement because the analyzer simply reproduces the entire spectrum of the
incoming waveform. However, the spectral content of the held samples without the envelope weighting is required for
proper measurement of the sample’s linearity, as would be measured by a downstream A/D converter that samples
a time instant in the held waveform. Either the impact of the response envelope must be corrected in the data or a
measurement method must be used that heterodynes the relevant nonlinear harmonic products to low frequencies
to avoid significant envelope response weighting. This latter method is referred to as the low frequency beat-product
technique.
The low frequency beat-product technique is commonly used for high-speed T/H linearity measurements, although
the measurement does impose restrictions on the specific input signal and clock frequencies that can be used. For
example, with a clock frequency of 512.5 MHz, a single tone input at 995 MHz beats with the 2nd harmonic of the
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