OPA686 Datasheet, PDF (9/16 Pages) Burr-Brown (TI) – Wideband, Low Noise, Voltage Feedback OPERATIONAL AMPLIFIER

transimpedance amplifier. One transimpedance design ex-
ample is shown on the front page of the data sheet. Designs
that require high bandwidth from a large area (high capaci-
tance) detector with relatively low transimpedance gain will
benefit from the low input voltage noise offered by the
OPA686. This input voltage noise will be peaked up over
frequency at the output by the diode source capacitance, and
can, in many cases, become the limiting factor to input
sensitivity. The key elements of the design are the expected
diode capacitance (C_D) with the reverse bias voltage (–V_B)
applied, the desired transimpedance gain, R_F, and the GBP
of the OPA686 (1600MHz). Figure 3 shows a design using
a 50pF source capacitance diode and a 10kΩ transimpedance
gain. With these three variables set (and including the
parasitic input capacitance for the OPA686 added to C_D), the
feedback capacitor value (C_F) may be set to control the
frequency response.
+5V
Supply Decoupling
Not Shown
OPA686
V_O = I_D R_F
R_F
10kΩ
λ
I_D
C_D
50pF
C_F
–5V _0.8pF
–V_B
FIGURE 3. Wideband, Low Noise, Transimpendance
Amplifier.
To achieve a maximally flat 2nd-order Butterworth fre-
quency response, the feedback pole should be set to:
1/(2πR_FC_F) = √(GBP/(4πR_FC_D))
Adding the common-mode and differential mode input ca-
pacitance (1.0 + 2.0)pF to the 50pF diode source capacitance
of Figure 3, and targeting a 10kΩ transimpedance gain using
the 1600MHz GBP for the OPA686, will require a feedback
pole set to 15.5MHz. This will require a total feedback
capacitance of 1.0pF. Typical surface-mount resistors have
a parasitic capacitance of 0.2pF, leaving the required 0.8pF
value shown in Figure 3 to get the required feedback pole.
This will give an approximate –3dB bandwidth equal to:
f_–3dB = √(GBP/2πR_FC_D)Hz
The example of Figure 3 will give approximately 23MHz
flat bandwidth using the 0.8pF feedback compensation.
If the total output noise is bandlimited to a frequency less
than the feedback pole frequency, a very simple expression
for the equivalent input noise current can be derived as:
Where:
I_EQ = Equivalent input noise current if the output noise is
bandlimited to F < 1/(2πR_FC_D)
I_N = Input current noise for the op amp inverting input
E_N = Input voltage noise for the op amp
C_D = Diode capacitance
F = Bandlimiting frequency in Hz (usually a post filter
prior to further signal processing)
Evaluating this expression up to the feedback pole frequency
at 15.5MHz for the circuit of Figure 3, gives an equivalent
input noise current of 6.4pA/√Hz. This is much higher than
the 1.8pA/√Hz for just the op amp itself. This result is being
dominated by the last term in the equivalent input noise
expression. It is essential in this case to use a low voltage
noise op amp. For example, if a slightly higher input noise
voltage, but otherwise identical op amp were used instead of
the OPA686 in this application (say 2.0nV/√Hz ), the total
input-referred current noise would increase to 9.5pA/√Hz.
LOW GAIN COMPENSATION FOR IMPROVED SFDR
Where a low gain is desired, and inverting operation is
acceptable, a new external compensation technique may be
used to retain the full slew rate and noise benefits of the
OPA686 while giving increased loop gain and the associ-
ated improvement in distortion offered by the decompen-
sated architecture. This technique shapes the loop gain for
good stability while giving an easily controlled second-
order low pass frequency response. Considering only the
noise gain (non-inverting signal gain) for the circuit of
Figure 4, the low frequency noise gain, (NG₁) will be set by
the resistor ratios while the high frequency noise gain (NG₂)
will be set by the capacitor ratios. The capacitor values set
both the transition frequencies and the high frequency noise
gain. If this noise gain, determined by NG₂ = 1+C_S/C_F, is set
to a value greater than the recommended minimum stable
gain for the op amp and the noise gain pole, set by 1/R_FC_F,
is placed correctly, a very well controlled, 2nd-order low
pass frequency response will result.
+5V
OPA686
V_O
R_G
250Ω
V_I
C_S
27pF
–5V
R_F
500Ω
C_F
2.9pF
( ) _{IEQ =}
I
2
N
+
4kT
R_F
+


E_N
R_F


2
+
E_N 2πC_DF ²
3
FIGURE 4. Broadband Low Gain Inverting External Com-
pensation.
®
9
OPA686