OPA2631 Datasheet, PDF (14/18 Pages) Burr-Brown (TI) – Dual, Low Power, Single-Supply OPERATIONAL AMPLIFIER

BANDWIDTH VERSUS GAIN: NON-INVERTING OPERATION
Voltage-feedback op amps exhibit decreasing closed-loop
bandwidth as the signal gain is increased. In theory, this
relationship is described by the Gain Bandwidth Product
(GBP) shown in the Specifications table. Ideally, dividing
GBP by the non-inverting signal gain (also called the Noise
Gain, or NG) will predict the closed-loop bandwidth. In
practice, this only holds true when the phase margin ap-
proaches 90°, as it does in high-gain configurations. At low
gains (increased feedback factors), most amplifiers will
exhibit a more complex response with lower phase margin.
The OPA2631 is compensated to give a slightly peaked
response in a non-inverting gain of 2 (Figure 1). This results
in a typical gain of +2 bandwidth of 75MHz, far exceeding
that predicted by dividing the 68MHz GBP by 2. Increasing
the gain will cause the phase margin to approach 90° and the
bandwidth to more closely approach the predicted value of
(GBP/NG). At a gain of +10, the 7.6MHz bandwidth shown
in the Specifications table is close to that predicted using the
simple formula and the typical GBP.
The OPA2631 exhibits minimal bandwidth reduction going
to +3V single-supply operation as compared with +5V
supply. This is because the internal bias control circuitry
retains nearly constant quiescent current as the total supply
voltage between the supply pins is changed.
INVERTING AMPLIFIER OPERATION
Since the OPA2631 is a general-purpose, wideband voltage-
feedback op amp, all of the familiar op amp application
circuits are available to the designer. Figure 7 shows a typical
inverting configuration where the I/O impedances and signal
gain from Figure 1 are retained in an inverting circuit configu-
ration. Inverting operation is one of the more common
requirements and offers several performance benefits. The
inverting configuration shows improved slew rate and distor-
tion. It also biases the input at V_S/2 for the best headroom. The
output voltage can be independently moved with bias adjust-
ment resistors connected to the inverting input.
0.1µF
+5V
2R_T
1.50kΩ
2R_T
1.50kΩ
+
0.1µF
6.8µF
1/2
OPA2631
R_O
50Ω
50Ω Load
50Ω
R_G
Source ^{0.1µF 374Ω}
R_M
57.6Ω
R_F
750Ω
In the inverting configuration, three key design consider-
ation must be noted. The first is that the gain resistor (R_G)
becomes part of the signal channel input impedance. If input
impedance matching is desired (which is beneficial when-
ever the signal is coupled through a cable, twisted pair, long
PC board trace, or other transmission line conductor), R_G
may be set equal to the required termination value, and R_F
adjusted to give the desired gain. This is the simplest
approach and results in optimum bandwidth and noise per-
formance. However, at low inverting gains, the resultant
feedback resistor value can present a significant load to the
amplifier output. For an inverting gain of 2, setting R_G to
50Ω for input matching eliminates the need for R_M but
requires a 100Ω feedback resistor. This has the interesting
advantage of the noise gain becoming equal to 2 for a 50Ω
source impedance—the same as the non-inverting circuits
considered above. However, the amplifier output will now
see the 100Ω feedback resistor in parallel with the external
load. In general, the feedback resistor should be limited to
the 200Ω to 1.5kΩ range. In this case, it is preferable to
increase both the R_F and R_G values, as shown in Figure 7,
and then achieve the input matching impedance with a third
resistor (R_M) to ground. The total input impedance becomes
the parallel combination of R_G and R_M.
The second major consideration, touched on in the previous
paragraph, is that the signal source impedance becomes
part of the noise gain equation and hence influences the
bandwidth. For the example in Figure 7, the R_M value
combines in parallel with the external 50Ω source imped-
ance, yielding an effective driving impedance of
50Ω || 576Ω = 26.8Ω. This impedance is added in series
with R_G for calculating the noise gain. The resultant is 2.87
for Figure 7, as opposed to only 2 if R_M could be eliminated
as discussed above. The bandwidth will therefore be lower
for the gain of –2 circuit of Figure 7 (NG = +2.87) than for
the gain of +2 circuit of Figure 1.
The third important consideration in inverting amplifier
design is setting the bias current cancellation resistors on
the non-inverting input (a parallel combination of
R_T = 750Ω). If this resistor is set equal to the total DC
resistance looking out of the inverting node, the output DC
error, due to the input bias currents, will be reduced to
(input offset current) • R_F. The inverting input's bias
current flows through R_F because of the 0.1µF capacitor.
Thus, we need R_T = 750Ω = 1.50kΩ || 1.50kΩ. To reduce
the additional high-frequency noise introduced by this R_T
resistor, and power-supply feedthrough, it is bypassed
with a capacitor. If we had R_T < 400Ω, its noise contribu-
tion would be minimal. As a minimum, the OPA2631
requires an R_T value of 50Ω to damp out parasitic-induced
peaking—a direct short to ground on the non-inverting
input runs the risk of a very high-frequency instability in
the input stage.
FIGURE 7. Gain of –2 Example Circuit.
14
OPA2631
SBOS067A