OPA2684 Datasheet, PDF (22/33 Pages) Texas Instruments – MINIMAL BANDWIDTH CHANGE VERSUS GAIN, 170MHz BANDWIDTH AT G = +2

lower 3rd-harmonic component. Focusing then on the 2nd-
harmonic, increasing the load impedance improves distortion
directly. Remember that the total load includes the feedback
network—in the noninverting configuration (see Figure 1) this
is the sum of R_F + R_G, while in the inverting configuration it
is just R_F. Also, providing an additional supply decoupling
capacitor (0.1µF) between the supply pins (for bipolar opera-
tion) improves the 2nd-order distortion slightly (3dB to 6dB).
In most op amps, increasing the output voltage swing in-
creases harmonic distortion directly. A low-power part like
the OPA2684 includes quiescent boost circuits to provide the
full-power bandwidth shown in the Typical Characteristics.
These act to increase the bias in a very linear fashion only
when high slew rate or output power are required. This also
acts to actually reduce the distortion slightly at higher output
power levels. The Typical Characteristics show the 2nd-
harmonic holding constant from 500mVp-p to 5Vp-p outputs
while the 3rd-harmonics actually decrease with increasing
output power.
The OPA2684 has an extremely low 3rd-order harmonic
distortion, particularly for light loads and at lower frequen-
cies. This also gives low 2-tone, 3rd-order intermodulation
distortion as shown in the Typical Characteristics. Since the
OPA2684 includes internal power boost circuits to retain
good full-power performance at high frequencies and out-
puts, it does not show a classical 2-tone, 3rd-order
intermodulation intercept characteristic. Instead, it holds rela-
tively low and constant 3rd-order intermodulation spurious
levels over power. The Typical Characteristics show this
spurious level as a dBc below the carrier at fixed center
frequencies swept over single-tone power at a matched 50Ω
load. These spurious levels drop significantly (> 12dB) for
lighter loads than the 100Ω used in that plot. Converter inputs
for instance will see ≤ 82dBc 3rd-order spurious to 10MHz for
full-scale inputs. For even lower 3rd-order intermodulation
distortion to much higher frequencies, consider the OPA2691.
NOISE PERFORMANCE
Wideband current-feedback op amps generally have a higher
output noise than comparable voltage-feedback op amps.
The OPA2684 offers an excellent balance between voltage
and current noise terms to achieve low output noise in a low
power amplifier. The inverting current noise (17pA/√Hz) is
lower most other current-feedback op amps while the input
voltage noise (3.7nV/√Hz) is lower than any unity-gain stable,
comparable slew rate, voltage-feedback op amp. This low
input voltage noise was achieved at the price of higher
noninverting input current noise (9.4pA/√Hz). As long as the
AC source impedance looking out of the noninverting node is
less than 200Ω, this current noise will not contribute signifi-
cantly to the total output noise. The op amp input voltage
noise and the two input current noise terms combine to give
low output noise under a wide variety of operating conditions.
Figure 16 shows the op amp noise analysis model with all the
noise terms included. In this model, all noise terms are taken
to be noise voltage or current density terms in either nV/√Hz
or pA/√Hz.
E_NI
OPA681
E_O
R_S
I_BN
E_RS
√4kTR_S
4kT
R_G
R_F
√4kTR_F
R_G
I_BI
4kT = 1.6E –20J
at 290°K
FIGURE 16. Op Amp Noise Analysis Model.
The total output spot noise voltage can be computed as the
square root of the sum of all squared output noise voltage
contributors. Equation 3 shows the general form for the
output noise voltage using the terms shown in Figure 16.
(3)
( ) ( ) E_O = ^E_NI2 + I_BNR_S ² + 4kTR_S ^NG² + I_BIR_F ² + 4kTR_FNG
Dividing this expression by the noise gain (NG = (1 + R_F/R_G))
will give the equivalent input referred spot noise voltage at
the noninverting input as shown in Equation 4.
(4)
( ) E_N =
E_NI2
+
I_BNR_S
2
+ 4kTR_S
+

I_BIR_F
NG

2
+
4kTR_F
NG
Evaluating these two equations for the OPA2684 circuit and
component values (see Figure 1) will give a total output spot
noise voltage of 16.3nV/√Hz and a total equivalent input spot
noise voltage of 8.2 nV/√Hz. This total input referred spot
noise voltage is higher than the 3.7nV/√Hz specification for
the op amp voltage noise alone. This reflects the noise
added to the output by the inverting current noise times the
feedback resistor. As the gain is increased, this fixed output
noise power term contributes less to the total output noise
and the total input referred voltage noise given by Equation 4
will approach just the 3.7nV/√Hz of the op amp itself. For
example, going to a gain of +20 in the circuit of Figure 1,
adjusting only the gain resistor to 42.1Ω, will give a total input
referred noise of 3.9nV/√Hz. A more complete description of
op amp noise analysis can be found in TI application note
AB-103, “Noise Analysis for High-Speed Op Amps”
(SBOA066), located at www.ti.com.
DC ACCURACY AND OFFSET CONTROL
A current-feedback op amp like the OPA2684 provides
exceptional bandwidth in high gains, giving fast pulse settling
but only moderate DC accuracy. The Electrical Characteris-
tics show an input offset voltage comparable to high slew
rate voltage-feedback amplifiers. The two input bias currents,
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OPA2684
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