NCV8852 Datasheet, PDF (10/12 Pages) ON Semiconductor – Automotive Grade Non-Synchronous Buck Controller

NCV8852
(2) Switching Frequency Selection
Selecting the switching frequency is a trade−off between
component size and power losses. Operation at higher
switching frequencies allows the use of smaller inductor and
capacitor values to achieve the same inductor current ripple
and output voltage ripple. However, increasing the
frequency increases the switching losses of the MOSFETs,
leading to decreased efficiency, especially noticeable at light
loads.
Typically, the switching frequency is selected to avoid
interfering with signals of known frequencies. The graph in
Figure 16, below, shows the required resistance to program
the frequency. From 200 kHz to 500 kHz, the following
formula is accurate to within 3% of the expected value:
R_OSC
+
2859
F_SW * 170
where:
F_SW: desired switching frequency [kHz]
R_OSC: resistor from ROSC pin to GND [kW]
110
100
90
80
70
60
50
40
30
20
10
0
100
200
300
400
500
600
F_SW (kHz)
Figure 16. Frequency vs. ROSC
(3) Current Sensor Selection
Current sensing for peak current mode control relies on
the inductor current signal. This is translated into a voltage
via a current sense resistor, which is then measured
differentially by the current sense amplifier, generating a
single−ended output to use as a signal. The easiest means of
implementing this transresistance is through the use of a
sense resistor in series with the source of the MOSFET and
VIN. A sense resistor should be calculated as follows:
R_SNS
+
V_CL
I_CL
where:
R_SNS: sense resistor [W]
V_CL: current limit threshold voltage [V]
I_CL: desired cycle−by−cycle current limit [A]
(4) MOSFET Selection
The NCV8852 has been designed to work with a
P−channel MOSFET in a non−synchronous buck
configuration. The MOSFET needs to be capable of
handling the maximum allowable current in the system, I_CL.
Keep in mind that, depending on your minimum V_IN signal,
it is possible to achieve 100% duty cycle. The power
dissipated through the MOSFET during conduction is as
follows:
P_MOS,on + I_CL ² @ D_MAX @ r_DS,on
where:
P_MOS,on: power through MOSFET [W]
I_CL: cycle−by−cycle current limit [A]
r_DS,on: on−resistance of the MOSFET [W]
To calculate the switching losses through the MOSFET, use
the following equation:
P_MOS,sw
+
1
2
V_IN
@
I_OUT
@
ǒ_ton
)
t_off^Ǔ
@
F_SW
t_on
+
t_off
+
Q_Gate
I_drv
where:
P_{MOS, sw}: MOSFET switching losses [W]
t_on: time to turn on the MOSFET [s]
t_off: time to turn off the MOSFET [s]
Q_Gate: gate charge [C]
I_drv: gate drive current [A]
(5) Diode Selection
The diode must be chosen according to its maximum
current and voltage ratings, and to thermal considerations.
The maximum reverse voltage the diode sees is the
maximum input voltage (with some margin in case of
ringing on the switch node). The maximum forward current
is the peak current limit of the NCV8852, or 150% of I_CL.
(6) Output Inductor Selection
Both mechanical and electrical considerations influence
the selection of an output inductor. From a mechanical
perspective, smaller inductor values generally correspond to
smaller physical size. Since the inductor is often one of the
largest components in the power supply, a minimum
inductor value is particularly important in space−
constrained applications. From an electrical perspective, an
inductor is chosen for a set amount of current ripple and to
assure adequate transient response.
The output inductor controls the current ripple that occurs
over a switching period. A high current ripple will result in
excessive power loss and ripple current requirements. A low
current ripple will result in a poor control signal and a slow
current slew rate in the event of a load transient. A good
starting point for peak−to−peak ripple is around 10% of the
inductor current.To choose the inductor value based on the
peak−to−peak ripple current, use the following equation:
i_L
+
V_OUT @ (1 * D_MIN)
L @ F_SW
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