MIC2169A Datasheet, PDF (8/17 Pages) Micrel Semiconductor – 500kHz PWM Synchronous Buck Control IC

MIC2169A
low-side switches. For applications where V_IN < 5V, the internal
V_DD regulator operates in dropout mode, and it is necessary
that the power MOSFETs used are sub-logic level and are in
full conduction mode for V_GS of 2.5V. For applications when
V_IN > 5V; logic-level MOSFETs, whose operation is specified
at V_GS = 4.5V must be used.
It is important to note the on-resistance of a MOSFET
increases with increasing temperature. A 75°C rise in junc-
tion temperature will increase the channel resistance of the
MOSFET by 50% to 75% of the resistance specified at 25°C.
This change in resistance must be accounted for when cal-
culating MOSFET power dissipation and in calculating the
value of current-sense (CS) resistor. Total gate charge is the
charge required to turn the MOSFET on and off under speci-
fied operating conditions (V_DS and V_GS). The gate charge
is supplied by the MIC2169A gate-drive circuit. At 500kHz
switching frequency and above, the gate charge can be a
significant source of power dissipation in the MIC2169A. At
low output load, this power dissipation is noticeable as a
reduction in efficiency. The average current required to drive
the high-side MOSFET is:
I_{G[high-side](avg)}  Q_G  f_S
where:
I_{G[high-side](avg)} = average high-side MOSFET gate
current.
Q_G = total gate charge for the high-side MOSFET taken from
manufacturer’s data sheet for V_GS = 5V.
The low-side MOSFET is turned on and off at V_DS = 0 because
the freewheeling diode is conducting during this time. The
switching loss for the low-side MOSFET is usually negligible.
Also, the gate-drive current for the low-side MOSFET is
more accurately calculated using CISS at V_DS = 0 instead
of gate charge.
For the low-side MOSFET:
I_{G[low-side](avg)}  C_ISS  V_GS  f_S
Since the current from the gate drive comes from the input volt-
age, the power dissipated in the MIC2169A due to gate drive is:
  ^P_GATEDRIVE ^{ V}_IN ^I_{G[high-side](avg)} ^{ I}_{G[low-side](avg)}
A convenient figure of merit for switching MOSFETs is the on
resistance times the total gate charge R_DS(ON) × Q_G. Lower
numbers translate into higher efficiency. Low gate-charge
logic-level MOSFETs are a good choice for use with the
MIC2169A.
Parameters that are important to MOSFET switch selection
are:
• Voltage rating
• On-resistance
• Total gate charge
The voltage ratings for the top and bottom MOSFET are
essentially equal to the input voltage. A safety factor of 20%
should be added to the V_DS(max) of the MOSFETs to account
for voltage spikes due to circuit parasitics.
The power dissipated in the switching transistor is the sum
Micrel
of the conduction losses during the on-time (P_CONDUCTION)
and the switching losses that occur during the period of time
when the MOSFETs turn on and off (P_AC).
P_SW  P_CONDUCTION  P_AC
where:
P_CONDUCTION  I_SW(rms)2  R_SW
P_AC  P_AC(off)  P_AC(on)
R_SW = on-resistance of the MOSFET switch
D

duty
cycle


V_O
V_IN


Making the assumption the turn-on and turn-off transition times
are equal; the transition times can be approximated by:
t_T

C_ISS

V_GS  C_OSS
I_G

V_IN
where:
C_ISS and C_OSS are measured at V_DS = 0
I_G = gate-drive current (1A for the MIC2169A)
The total high-side MOSFET switching loss is:
P_AC  (V_IN V_D )  I_PK  t_T  f_S
where:
t_T = switching transition time (typically 20ns to 50ns)
V_D = freewheeling diode drop, typically 0.5V
f_S it the switching frequency, nominally 500kHz
The low-side MOSFET switching losses are negligible and
can be ignored for these calculations.
Inductor Selection
Values for inductance, peak, and RMS currents are required
to select the output inductor. The input and output voltages
and the inductance value determine the peak-to-peak induc-
tor ripple current. Generally, higher inductance values are
used with higher input voltages. Larger peak-to-peak ripple
currents will increase the power dissipation in the inductor
and MOSFETs. Larger output ripple currents will also require
more output capacitance to smooth out the larger ripple cur-
rent. Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more expensive
inductor. A good compromise between size, loss and cost is
to set the inductor ripple current to be equal to 20% of the
maximum output current. The inductance value is calculated
by the equation below.
L  ^{VOUT (VINmax  VOUT )}
V_INmax  f_S  0.2  I_OUT max
where:
f_S = switching frequency, 500kHz
0.2 = ratio of AC ripple current to DC output current
V_IN(max) = maximum input voltage
The peak-to-peak inductor current (AC ripple current) is:
M9999-111803
8
June 2005