OPA2674 Datasheet, PDF (21/30 Pages) Burr-Brown (TI) – Dual Wideband, High Output Current Operational Amplifier with Current Limit

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supplies. Evaluating the denominator of Equation 15
(which is the feedback transimpedance) gives an optimal
target of 490Ω. As the signal gain changes, the
contribution of the NG × R_I term in the feedback
transimpedance changes, but the total can be held
constant by adjusting R_F. Equation 16 gives an
approximate equation for optimum R_F over signal gain:
R_F + 490 * NG R_I
(16)
As the desired signal gain increases, this equation
eventually suggests a negative R_F. A somewhat subjective
limit to this adjustment can also be set by holding R_G to a
minimum value of 20Ω. Lower values load both the buffer
stage at the input and the output stage if R_F gets too
lowactually decreasing the bandwidth. Figure 11 shows
the recommended R_F versus NG for both ±6V and a single
+5V operation. The values for R_F versus gain shown here
are approximately equal to the values used to generate the
Typical Characteristics. They differ in that the optimized
values used in the Typical Characteristics are also
correcting for board parasitic not considered in the
simplified analysis leading to Equation 16. The values
shown in Figure 11 give a good starting point for designs
where bandwidth optimization is desired.
600
500
+5V
400
300
±6V
R_G = 20Ω
200
0
5
10
15
20
25
Noise Gain
Figure 11. Feedback Resistor vs Noise Gain
The total impedance going into the inverting input may be
used to adjust the closed-loop signal bandwidth. Inserting
a series resistor between the inverting input and the
summing junction increases the feedback impedance (the
denominator of Equation 15), decreasing the bandwidth.
The internal buffer output impedance for the OPA2674 is
slightly influenced by the source impedance coming from
of the noninverting input terminal. High-source resistors
also have the effect of increasing R_I, decreasing the
bandwidth. For those single-supply applications that
develop a midpoint bias at the noninverting input through
high valued resistors, the decoupling capacitor is essential
for power-supply ripple rejection, noninverting input noise
current shunting, and to minimize the high-frequency
value for R_I in Figure 10.
OPA2674
SBOS270 − AUGUST 2003
INVERTING AMPLIFIER OPERATION
As the OPA2674 is a general-purpose, wideband
current-feedback op amp, most of the familiar op amp
application circuits are available to the designer. Those
dual op amp applications that require considerable
flexibility in the feedback element (for example,
integrators, transimpedance, and some filters) should
consider a unity-gain stable, voltage-feedback amplifier
such as the OPA2822, because the feedback resistor is
the compensation element for a current-feedback op amp.
Wideband inverting operation (and especially summing) is
particularly suited to the OPA2674. Figure 12 shows a
typical inverting configuration where the I/O impedances
and signal gain from Figure 1 are retained in an inverting
circuit configuration.
50Ω
Source
V_I
R_G
97.6Ω
R_M
102Ω
+6V
Power−supply
decoupling not
shown.
1/2
V_O
O PA 267 4
50Ω Load
50Ω
R_F
392Ω
−6V
Figure 12. Inverting Gain of −4 with Impedance
Matching
In the inverting configuration, two key design
considerations must be noted. First, the gain resistor (R_G)
becomes part of the signal source input impedance. If
input impedance matching is desired (which is beneficial
whenever the signal is coupled through a cable, twisted
pair, long PC board trace, or other transmission line
conductor), it is normally necessary to add an additional
matching resistor to ground. R_G, by itself, normally is not
set to the required input impedance since its value, along
with the desired gain, will determine an R_F, which may be
nonoptimal from a frequency response standpoint. The
total input impedance for the source becomes the parallel
combination of R_G and R_M.
The second major consideration is that the signal source
impedance becomes part of the noise gain equation and
has a slight effect on the bandwidth through Equation 15.
The values shown in Figure 12 have accounted for this by
slightly decreasing R_F (from the optimum values) to
reoptimize the bandwidth for the noise gain of Figure 12
(NG = 3.98). In the example of Figure 12, the R_M value
combines in parallel with the external 50Ω source
impedance, yielding an effective driving impedance of
50Ω || 102Ω = 33.5Ω. This impedance is added in series
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