AD8309_15 Datasheet, PDF (14/20 Pages) Analog Devices – 5 MHz–500 MHz 100 dB Demodulating Logarithmic Amplifier with Limiter Output

AD8309
load current, which may be large, the value of R2 should take
this into account.
The four pins labeled PADL tie down directly to the metallic
lead frame, and are thus connected to the back of the chip. The
process on which the AD8309 is fabricated uses a bonded-wafer
technique to provide a silicon-on-insulator isolation, and there is
no junction or other dc path from the back side to the circuitry
on the surface. These paddle pins must be connected directly to
the ground plane using the shortest possible lead lengths to
minimize inductance.
Basic Connections
Figure 30 shows the connections required for most applications.
The inputs are ac-coupled by C1 and C2, which normally
should have the same value, say, C_O. The coupling time con-
stant is R_OC_O /2, where R_O = R_S + R_IN, thus forming a high pass
corner with a 3 dB attenuation at f_HP = 1/(π R_T C_C ). In high-
frequency applications, f_HP should be chosen as large as pos-
sible, to minimize the coupling of unwanted signals. On the
other hand, in low frequency applications, a simple RC network
forming a low-pass filter should be added at the input for the
same reason.
2.5
100MHz
50MHz
200MHz
5MHz
2.0
1.5
1.0
0.5
0
–100 –80
–60
–40 –20
0
20
40
INPUT LEVEL – dBm Re 50⍀
Figure 31. RSSI Output vs. Input Level at T_A = +25°C, for
Frequencies of 5 MHz, 50 MHz, 100 MHz and 200 MHz
5
4
3
5MHz
DYNAMIC RANGE 1dB 3dB
5MHz
85 93
50MHz
91 99
100MHz
97 103
200MHz
96 102
R1
10⍀
SEE TEXT FOR MORE
ABOUT DECOUPLING
0.1␮F
1 COM2
2 VPS1
VLOG 16
VPS2 15
R2
10⍀
0.1␮F
V_S
RSSI
C1
SIGNAL
INPUTS
C2
R_T
52.3⍀
4.7nH
ENABLE
FOR BROADBAND 50⍀
TERMINATION TO 1GHz
3 PADL
PADL 14
AD8309
4 INHI
LMHI 13
5 INLO
LMLO 12
6 PADL
PADL 11
7 COM1
8 ENBL
FLTR 10 NC
R_LIM
LMDR 9
NC = NO CONNECT
LMHI
R_LOAD
R_LOAD
LMLO
2
1
50MHz
0
–1
100MHz ^200MHz
–2
–3
–4
–5
–90 –80 –70 –60 –50 –40 –30 –20 –10 0
INPUT LEVEL – dBm Re 50⍀
10 20 30
Figure 32. Log Linearity vs. Input Level at T_A = +25°C, for
Frequencies of 5 MHz, 50 MHz, 100 MHz and 200 MHz
Figure 30. Basic Connections
Where it is necessary to terminate the source at a low imped-
ance, the resistor R_T should be added, with allowance for the
shunting effect of the 1 kΩ input resistance (R_IN) of the AD8309.
For example, to terminate a 50 Ω source, a 52.3 Ω␣ resistor
should be used for signal frequencies up to about 50 MHz. The
termination means may be placed either at the input or at the
log amp side of the coupling capacitors. In the former case
smaller capacitors can be used for a given frequency range; in
the latter case, the dc resistance is lowered directly at the log
amp inputs, which helps to keep offsets to a minimum. At
higher frequencies, the reactance of the 2.5 pF input capaci-
tance must be accounted for. A 4.7 nH inductor in series with
the 52.3 Ω termination resistor provides an essentially flat 50 Ω
input impedance to 1 GHz. An impedance-transforming net-
work is preferably used to provide a 50 Ω interface, since this
also introduces a balanced voltage gain of typically 13 dB and
the AD8309 has a very high capacity for large input voltages.
Figure 31 shows the output versus the input level, with the axis
marked in dBm (correct only when terminated in 50 Ω), for sine
inputs at 5 MHz, 50 MHz, 100 MHz and 200 MHz. Figure 32
shows the typical logarithmic linearity (law conformance) under
the same conditions.
Input Matching
Where either a higher sensitivity or a better high frequency
match is required, an input matching network is valuable. Using
a flux-coupled transformer to achieve the impedance transfor-
mation also eliminates the need for coupling capacitors, lowers
any dc offset voltages generated directly at the input, and use-
fully balances the drives to INHI and INLO, permitting full
utilization of the unusually large input voltage capacity of the
AD8309.
The choice of turns ratio will depend somewhat on the fre-
quency. At frequencies below 30 MHz, the reactance of the
input capacitance is much higher than the real part of the input
impedance. In this frequency range, a turns ratio of 2:9 will
lower the effective input impedance to 50 Ω while raising the
input voltage by 13 dB. However, this does not lower the effect
of the short circuit noise voltage by the same factor, since there
will be a contribution from the input noise current. Thus, the
total noise will be reduced by a smaller factor. The intercept at
the primary input will be lowered to –120 dBV (–107 dBm).
Impedance matching and drive balancing using a flux-coupled
transformer is useful whenever broadband coupling is required.
However, this may not always be convenient. At high frequen-
cies, it will often be preferable to use a narrow-band matching
network, as shown in Figure 33, which has several advantages.
First, the same voltage gain can be achieved, providing increased
–14–
REV. B